LM2575T-12 - ON Semiconductor

 Semiconductor Components Industries, LLC, 2002

January, 2002 – Rev. 4 1Publication Order Number:

LM2575/D

LM2575

1.0 A, Adjustable Output

Voltage, Step-Down

Switching Regulator

The LM2575 series of regulators are monolithic integrated circuits

ideally suited for easy and convenient design of a step–down

switching regulator (buck converter). All circuits of this series are

capable of driving a 1.0 A load with excellent line and load regulation.

These devices are available in fixed output voltages of 3.3 V, 5.0 V,

12 V, 15 V, and an adjustable output version.

These regulators were designed to minimize the number of external

components to simplify the power supply design. Standard series of

inductors optimized for use with the LM2575 are offered by several

different inductor manufacturers.

Since the LM2575 converter is a switch–mode power supply, its

efficiency is significantly higher in comparison with popular

three–terminal linear regulators, especially with higher input voltages.

In many cases, the power dissipated by the LM2575 regulator is so

low, that no heatsink is required or its size could be reduced

dramatically.

The LM2575 features include a guaranteed ±4% tolerance on output

voltage within specified input voltages and output load conditions, and

±10% on the oscillator frequency (±2% over 0°C to 125°C). External

shutdown is included, featuring 80 µA typical standby current. The

output switch includes cycle–by–cycle current limiting, as well as

thermal shutdown for full protection under fault conditions.

Features

•3.3 V, 5.0 V, 12 V, 15 V, and Adjustable Output Versions

•Adjustable Version Output Voltage Range of 1.23 V to 37 V ±4%

Maximum Over Line and Load Conditions

•Guaranteed 1.0 A Output Current

•Wide Input Voltage Range: 4.75 V to 40 V

•Requires Only 4 External Components

•52 kHz Fixed Frequency Internal Oscillator

•TTL Shutdown Capability, Low Power Standby Mode

•High Efficiency

•Uses Readily Available Standard Inductors

•Thermal Shutdown and Current Limit Protection

•Moisture Sensitivity Level (MSL) Equals 1

Applications

•Simple and High–Efficiency Step–Down (Buck) Regulators

•Efficient Pre–Regulator for Linear Regulators

•On–Card Switching Regulators

•Positive to Negative Converters (Buck–Boost)

•Negative Step–Up Converters

•Power Supply for Battery Chargers

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See detailed ordering and shipping information in the package

dimensions section on page 24 of this data sheet.

ORDERING INFORMATION

TO–220

TV SUFFIX

CASE 314B

Heatsink surface connected to Pin 3

TO–220

T SUFFIX

CASE 314D

Pin 1. Vin

2. Output

3. Ground

4. Feedback

5. ON/OFF

D2PAK

D2T SUFFIX

CASE 936A

Heatsink surface (shown as terminal 6 in

case outline drawing) is connected to Pin 3

See general marking information in the device marking

section on page 24 of this data sheet.

DEVICE MARKING INFORMATION

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Figure 1. Block Diagram and Typical Application

7.0 V - 40 V

Unregulated

DC Input

330 µH

Gnd

+Vin

Cin

100 µF3ON

/OFF5

Output

Feedback

1N5819 Cout

330 µF

Typical Application (Fixed Output Voltage Versions)

Representative Block Diagram and Typical Application

Unregulated

DC Input +Vin

Cout

Feedback

Cin

1.0 k Output

Gnd

ON/OFF

Reset

Latch

Thermal

Shutdown

52 kHz

Oscillator

1.235 V

Band-Gap

Reference

Freq

Shift

18 kHz

Comparator

Fixed Gain

Error Amplifier

Current

Limit

Driver

1.0 Amp

Switch

ON/OFF

3.1 V Internal

Regulator

Regulated

Output

Vout

Load

Output

Voltage Versions

3.3 V

5.0 V

12 V

15 V

(Ω)

1.7 k

3.1 k

8.84 k

11.3 k

For adjustable version

R1 = open, R2 = 0 Ω

LM2575

5.0 V Regulated

Output 1.0 A Load

This device contains 162 active transistors.

ABSOLUTE MAXIMUM RATINGS (Absolute Maximum Ratings indicate limits beyond which damage to the device may occur.)

Rating Symbol Value Unit

Maximum Supply Voltage Vin 45 V

ON/OFF Pin Input Voltage ––0.3 V ≤ V ≤ +Vin V

Output Voltage to Ground (Steady–State) – –1.0 V

Power Dissipation

Case 314B and 314D (TO–220, 5–Lead) PDInternally Limited W

Thermal Resistance, Junction–to–Ambient RθJA 65 °C/W

Thermal Resistance, Junction–to–Case RθJC 5.0 °C/W

Case 936A (D2PAK) PDInternally Limited W

Thermal Resistance, Junction–to–Ambient (Figure 34) RθJA 70 °C/W

Thermal Resistance, Junction–to–Case RθJC 5.0 °C/W

Storage Temperature Range Tstg –65 to +150 °C

Minimum ESD Rating (Human Body Model: C = 100 pF, R = 1.5 kΩ) – 3.0 kV

Lead Temperature (Soldering, 10 s) – 260 °C

Maximum Junction Temperature TJ150 °C

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OPERATING RATINGS (Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee

specific performance limits. For guaranteed specifications and test conditions, see the Electrical Characteristics.)

Rating Symbol Value Unit

Operating Junction Temperature Range TJ–40 to +125 °C

Supply Voltage Vin 40 V

SYSTEM PARAMETERS ([Note 1] Test Circuit Figure 14)

ELECTRICAL CHARACTERISTICS (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and Adjustable version, Vin = 25 V

for the 12 V version, and Vin = 30 V for the 15 V version. ILoad = 200 mA. For typical values TJ = 25°C, for min/max values TJ is the

operating junction temperature range that applies [Note 2], unless otherwise noted.)

Characteristics Symbol Min Typ Max Unit

LM2575–3.3 ([Note 1] Test Circuit Figure 14)

Output Voltage (Vin = 12 V, ILoad = 0.2 A, TJ = 25°C) Vout 3.234 3.3 3.366 V

Output Voltage (4.75 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) Vout V

TJ = 25°C 3.168 3.3 3.432

TJ = –40 to +125°C 3.135 – 3.465

Efficiency (Vin = 12 V, ILoad = 1.0 A) η– 75 – %

LM2575–5 ([Note 1] Test Circuit Figure 14)

Output Voltage (Vin = 12 V, ILoad = 0.2 A, TJ = 25°C) Vout 4.9 5.0 5.1 V

Output Voltage (8.0 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) Vout V

TJ = 25°C 4.8 5.0 5.2

TJ = –40 to +125°C 4.75 – 5.25

Efficiency (Vin = 12 V, ILoad = 1.0 A) η– 77 – %

LM2575–12 ([Note 1] Test Circuit Figure 14)

Output Voltage (Vin = 25 V, ILoad = 0.2 A, TJ = 25°C) Vout 11.76 12 12.24 V

Output Voltage (15 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) Vout V

TJ = 25°C 11.52 12 12.48

TJ = –40 to +125°C 11.4 – 12.6

Efficiency (Vin = 15V, ILoad = 1.0 A) η– 88 – %

LM2575–15 ([Note 1] Test Circuit Figure 14)

Output Voltage (Vin = 30 V, ILoad = 0.2 A, TJ = 25°C) Vout 14.7 15 15.3 V

Output Voltage (18 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) Vout V

TJ = 25°C 14.4 15 15.6

TJ = –40 to +125°C 14.25 – 15.75

Efficiency (Vin = 18 V, ILoad = 1.0 A) η– 88 – %

LM2575 ADJUSTABLE VERSION ([Note 1] Test Circuit Figure 14)

Feedback Voltage (Vin = 12 V, ILoad = 0.2 A, Vout = 5.0 V, TJ = 25°C) VFB 1.217 1.23 1.243 V

Feedback Voltage (8.0 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A, Vout = 5.0 V) VFB V

TJ = 25°C 1.193 1.23 1.267

TJ = –40 to +125°C 1.18 – 1.28

Efficiency (Vin = 12 V, ILoad = 1.0 A, Vout = 5.0 V) η– 77 – %

1. External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance.

When the LM2575 is used as shown in the

Figure 14

test circuit, system performance will be as shown in system parameters section

2. Tested junction temperature range for the LM2575: Tlow = –40°C Thigh = +125°C

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DEVICE PARAMETERS

ELECTRICAL CHARACTERISTICS (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and Adjustable version, Vin = 25 V

for the 12 V version, and Vin = 30 V for the 15 V version. ILoad = 200 mA. For typical values TJ = 25°C, for min/max values TJ is the

operating junction temperature range that applies [Note 2], unless otherwise noted.)

Characteristics Symbol Min Typ Max Unit

ALL OUTPUT VOLTAGE VERSIONS

Feedback Bias Current (Vout = 5.0 V [Adjustable Version Only]) IbnA

TJ = 25°C – 25 100

TJ = –40 to +125°C – – 200

Oscillator Frequency [Note 3] fosc kHz

TJ = 25°C – 52 –

TJ = 0 to +125°C 47 – 58

TJ = –40 to +125°C 42 – 63

Saturation Voltage (Iout = 1.0 A [Note 4]) Vsat V

TJ = 25°C – 1.0 1.2

TJ = –40 to +125°C – – 1.3

Max Duty Cycle (“on”) [Note 5] DC 94 98 – %

Current Limit (Peak Current [Notes 4 and 3]) ICL A

TJ = 25°C 1.7 2.3 3.0

TJ = –40 to +125°C 1.4 – 3.2

Output Leakage Current [Notes 6 and 7], TJ = 25°C ILmA

Output = 0 V – 0.8 2.0

Output = –1.0 V – 6.0 20

Quiescent Current [Note 6] IQmA

TJ = 25°C – 5.0 9.0

TJ = –40 to +125°C – – 11

Standby Quiescent Current (ON/OFF Pin = 5.0 V (“off”)) Istby µA

TJ = 25°C – 80 200

TJ = –40 to +125°C – – 400

ON/OFF Pin Logic Input Level (Test Circuit Figure 14) V

Vout = 0 V VIH

TJ = 25°C 2.2 1.4 –

TJ = –40 to +125°C 2.4 – –

Vout = Nominal Output Voltage VIL

TJ = 25°C – 1.2 1.0

TJ = –40 to +125°C – – 0.8

ON/OFF Pin Input Current (Test Circuit Figure 14) µA

ON/OFF Pin = 5.0 V (“off”), TJ = 25°C IIH – 15 30

ON/OFF Pin = 0 V (“on”), TJ = 25°C IIL – 0 5.0

3. The oscillator frequency reduces to approximately 18 kHz in the event of an output short or an overload which causes the regulated output

voltage to drop approximately 40% from the nominal output voltage. This self protection feature lowers the average dissipation of the IC by

lowering the minimum duty cycle from 5% down to approximately 2%.

4. Output (Pin 2) sourcing current. No diode, inductor or capacitor connected to output pin.

5. Feedback (Pin 4) removed from output and connected to 0 V.

6. Feedback (Pin 4) removed from output and connected to +12 V for the Adjustable, 3.3 V, and 5.0 V versions, and +25 V for the 12 V and

15 V versions, to force the output transistor “off”.

7. Vin = 40 V.

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TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 14)

out, OUTPUT VOLTAGE CHANGE (%)

-50

3.0

-50

2.0

1.2

-50

IQ, QUIESCENT CURRENT (mA)

Vin, INPUT VOLTAGE (V)

IO, OUTPUT CURRENT (A)

TJ, JUNCTION TEMPERATURE (°C)

Vin, INPUT VOLTAGE (V)

INPUT-OUTPUT DIFFERENTIAL (V)

TJ, JUNCTION TEMPERATURE (°C)

Vsat , SATURATION VOLTAGE (V)

SWITCH CURRENT (A)

out, OUTPUT VOLTAGE CHANGE (%)

Figure 2. Normalized Output Voltage

TJ, JUNCTION TEMPERATURE (°C)

Figure 3. Line Regulation

Vin = 20 V

ILoad = 200 mA

Normalized at

TJ = 25°C

Figure 4. Switch Saturation Voltage Figure 5. Current Limit

Figure 6. Dropout Voltage Figure 7. Quiescent Current

ILoad = 200 mA

TJ = 25°C

3.3 V, 5.0 V and Adj

12 V and 15 V

25°C

Vin = 25 V

Vout = 5.0 V

Measured at

Ground Pin

TJ = 25°C

ILoad = 200 mA

ILoad = 1.0 A

∆Vout = 5%

Rind = 0.2 Ω

125°C

-40°C

5.0-25 100 201525 257550 3530 40100 125

0.8

0.4

-0.2

-0.4

0.6

0.2

1.00.6

0.2

-0.2

-0.6

2.5

1.5

0.5

2.0

1.0

6.0

4.0

8.0

1.1

0.9

0.7

0.5

1.0

0.8

0.6

1.2

0.8

0.4

1.0

0.6

1.8

1.4

1.6

0.4 -250.1 00.2 250.3 500.4 750.5 1000.6 1250.7

5.0-25 100 1525 2050 2575 30100 35125

0.8 0.9 1.0

ILoad = 200 mA

ILoad = 1.0 A

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OUTPUT

VOLTAGE

(PIN 2)

OUTPUT

CURRENT

(PIN 2)

INDUCTOR

OUTPUT

RIPPLE

VOLTAGE

Vout, OUTPUT VOLTAGE

Istby, STANDBY QUIESCENT CURRENT ( A)µ

100

-50

10 V

-50

100 µs/DIV

IFB, FEEDBACK PIN CURRENT (nA)

TJ, JUNCTION TEMPERATURE (°C)

5.0 µs/DIV

NORMALIZED FREQUENCY (%)

TJ, JUNCTION TEMPERATURE (°C)

Istby, STANDBY QUIESCENT CURRENT ( A)µ

Figure 8. Standby Quiescent Current

Vin, INPUT VOLTAGE (V)

Figure 9. Standby Quiescent Current

Figure 10. Oscillator Frequency Figure 11. Feedback Pin Current

Figure 12. Switching Waveforms Figure 13. Load Transient Response

Vin = 12 V

VON/OFF = 5.0 V

TJ = 25°C

-1001.0 A

1.0

2.0

0.5

1.0 A

120

100

0.5 A

-2.0

100

-40

-4.0

20 mV

-8.0

-10

120

-6.0

/DIV

ILoad, LOAD CURRENT (A)

-20

-25

5.0

100

125

4035

Vin = 12 V

Normalized at 25°C

Adjustable

Version Only

CHANGE (mV)

CURRENT

LM2575

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Figure 14. Typical Test Circuit

1N5819

330 µH

Output

Feedback

Cout

330 µF

/16 V

Cin

100 µF/50 V

LM2575–5

53ON

/OFFGnd

Vin

Load

Vout

Regulated

Output

Vin

Unregulated

DC Input

8.0 V - 40 V

1N5819

330 µH

Output

Feedback

Cout

330 µF

/16 V

Cin

100 µF/50 V

LM2575

Adjustable

53ON

/OFFGnd

Vin

Load

Vout

Regulated

Output

Unregulated

DC Input

8.0 V - 40 V

5.0 Output Voltage Versions

Adjustable Output Voltage Versions

Vout Vref1R2

R1

R2 R1Vout

Vref

1

Where Vref = 1.23 V, R1

between 1.0 kΩ and 5.0 kΩ

PCB LAYOUT GUIDELINES

As in any switching regulator, the layout of the printed

circuit board is very important. Rapidly switching currents

associated with wiring inductance, stray capacitance and

parasitic inductance of the printed circuit board traces can

generate voltage transients which can generate

electromagnetic interferences (EMI) and affect the desired

operation. As indicated in the Figure 14, to minimize

inductance and ground loops, the length of the leads

indicated by heavy lines should be kept as short as possible.

For best results, single–point grounding (as indicated) or

ground plane construction should be used.

On the other hand, the PCB area connected to the Pin 2

(emitter of the internal switch) of the LM2575 should be

kept to a minimum in order to minimize coupling to sensitive

circuitry.

Another sensitive part of the circuit is the feedback. It is

important to keep the sensitive feedback wiring short. To

assure this, physically locate the programming resistors near

to the regulator, when using the adjustable version of the

LM2575 regulator.

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PIN FUNCTION DESCRIPTION

Pin Symbol Description (Refer to Figure 1)

1 Vin This pin is the positive input supply for the LM2575 step–down switching regulator. In order to minimize

voltage transients and to supply the switching currents needed by the regulator, a suitable input bypass

capacitor must be present (Cin in

Figure 1).

2 Output This is the emitter of the internal switch. The saturation voltage Vsat of this output switch is typically 1.0 V.

It should be kept in mind that the PCB area connected to this pin should be kept to a minimum in order to

minimize coupling to sensitive circuitry.

3 Gnd Circuit ground pin. See the information about the printed circuit board layout.

4 Feedback This pin senses regulated output voltage to complete the feedback loop. The signal is divided by the

internal resistor divider network R2, R1 and applied to the non–inverting input of the internal error amplifier.

In the Adjustable version of the LM2575 switching regulator this pin is the direct input of the error amplifier

and the resistor network R2, R1 is connected externally to allow programming of the output voltage.

5 ON/OFF It allows the switching regulator circuit to be shut down using logic level signals, thus dropping the total

input supply current to approximately 80 µA. The input threshold voltage is typically 1.4 V. Applying a

voltage above this value (up to +Vin) shuts the regulator off. If the voltage applied to this pin is lower than

1.4 V or if this pin is connected to ground, the regulator will be in the “on” condition.

DESIGN PROCEDURE

Buck Converter Basics

The LM2575 is a “Buck” or Step–Down Converter which

is the most elementary forward–mode converter. Its basic

schematic can be seen in Figure 15.

The operation of this regulator topology has two distinct

time periods. The first one occurs when the series switch is

on, the input voltage is connected to the input of the inductor.

The output of the inductor is the output voltage, and the

rectifier (or catch diode) is reverse biased. During this

period, since there is a constant voltage source connected

across the inductor, the inductor current begins to linearly

ramp upwards, as described by the following equation:

IL(on) Vin –V

outton

During this “on” period, energy is stored within the core

material in the form of magnetic flux. If the inductor is

properly designed, there is sufficient energy stored to carry

the requirements of the load during the “off” period.

Figure 15. Basic Buck Converter

Vin

Vout

RLoad

Cout

Power

Switch

The next period is the “off” period of the power switch.

When the power switch turns off, the voltage across the

inductor reverses its polarity and is clamped at one diode

voltage drop below ground by catch dioded. Current now

flows through the catch diode thus maintaining the load

current loop. This removes the stored energy from the

inductor. The inductor current during this time is:

IL(off) Vout –V

Dtoff

This period ends when the power switch is once again

turned on. Regulation of the converter is accomplished by

varying the duty cycle of the power switch. It is possible to

describe the duty cycle as follows:

dton

T, where T is the period of switching.

For the buck converter with ideal components, the duty

cycle can also be described as:

dVout

Vin

Figure 16 shows the buck converter idealized waveforms

of the catch diode voltage and the inductor current.

Power

Switch

Figure 16. Buck Converter Idealized Waveforms

Power

Switch

Off

Power

Switch

Off

Power

Switch

Power

Switch

Von(SW)

VD(FWD)

Time

ILoad(AV)

Imin

Ipk

Diode Diode

Power

Switch

Diode VoltageInductor Current

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Procedure

(Fixed Output Voltage Version) In order to simplify the switching regulator design, a step–by–step

design procedure and example is provided.

Procedure Example

Given Parameters:

Vout = Regulated Output Voltage (3.3 V, 5.0 V, 12 V or 15 V)

Vin(max) = Maximum DC Input Voltage

ILoad(max) = Maximum Load Current

Given Parameters:

Vout = 5.0 V

Vin(max) = 20 V

ILoad(max) = 0.8 A

1. Controller IC Selection

According to the required input voltage, output voltage and

current, select the appropriate type of the controller IC output

voltage version.

1. Controller IC Selection

According to the required input voltage, output voltage,

current polarity and current value, use the LM2575–5

controller IC

2. Input Capacitor Selection (Cin)

To prevent large voltage transients from appearing at the input

and for stable operation of the converter, an aluminium or

tantalum electrolytic bypass capacitor is needed between the

input pin +Vin and ground pin Gnd. This capacitor should be

located close to the IC using short leads. This capacitor should

have a low ESR (Equivalent Series Resistance) value.

2. Input Capacitor Selection (Cin)

A 47 µF, 25 V aluminium electrolytic capacitor located near

to the input and ground pins provides sufficient bypassing.

3. Catch Diode Selection (D1)

A.Since the diode maximum peak current exceeds the

regulator maximum load current the catch diode current

rating must be at least 1.2 times greater than the maximum

load current. For a robust design the diode should have a

current rating equal to the maximum current limit of the

LM2575 to be able to withstand a continuous output short

B.The reverse voltage rating of the diode should be at least

1.25 times the maximum input voltage.

3. Catch Diode Selection (D1)

A.For this example the current rating of the diode is 1.0 A.

B.Use a 30 V 1N5818 Schottky diode, or any of the

suggested fast recovery diodes shown in the Table 4.

4. Inductor Selection (L1)

A.According to the required working conditions, select the

correct inductor value using the selection guide from

Figures 17 to 21

B.From the appropriate inductor selection guide, identify the

inductance region intersected by the Maximum Input

Voltage line and the Maximum Load Current line. Each

region is identified by an inductance value and an inductor

code.

C.Select an appropriate inductor from the several different

manufacturers part numbers listed in Table 1 or Table 2.

When using Table 2 for selecting the right inductor the

designer must realize that the inductor current rating must

be higher than the maximum peak current flowing through

the inductor. This maximum peak current can be calculated

as follows

where ton is the “on” time of the power switch and

For additional information about the inductor, see the

inductor section in the “External Components” section of

this data sheet.

Ip(max) ILoad(max)Vin–Voutton

ton Vout

Vin x1

fosc

4. Inductor Selection (L1)

A.Use the inductor selection guide shown in Figures 17

to 21.

B.From the selection guide, the inductance area intersected

by the 20 V line and 0.8 A line is L330.

C.Inductor value required is 330 µH. From the Table 1 or

Table 2

choose an inductor from any of the listed

manufacturers.

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Procedure

(Fixed Output Voltage Version) (continued)In order to simplify the switching regulator design, a step–by–step

design procedure and example is provided.

Procedure Example

5. Output Capacitor Selection (Cout)

A.Since the LM2575 is a forward–mode switching regulator

with voltage mode control, its open loop 2–pole–2–zero

frequency characteristic has the dominant pole–pair

determined by the output capacitor and inductor values. For

stable operation and an acceptable ripple voltage,

(approximately 1% of the output voltage) a value between

100 µF and 470 µF is recommended.

B.Due to the fact that the higher voltage electrolytic capacitors

generally have lower ESR (Equivalent Series Resistance)

numbers, the output capacitor’s voltage rating should be at

least 1.5 times greater than the output voltage. For a 5.0 V

regulator, a rating at least 8V is appropriate, and a 10 V or

16 V rating is recommended.

5. Output Capacitor Selection (Cout)

A.Cout = 100 µF to 470 µF standard aluminium electrolytic.

B.Capacitor voltage rating = 16 V.

Procedure (Adjustable Output Version:

LM2575–Adj)

Procedure Example

Given Parameters:

Vout = Regulated Output Voltage

Vin(max) = Maximum DC Input Voltage

ILoad(max) = Maximum Load Current

Given Parameters:

Vout = 8.0 V

Vin(max) = 12 V

ILoad(max) = 1.0 A

1. Programming Output Voltage

To select the right programming resistor R1 and R2 value (see

Figure 14) use the following formula:

Resistor R1 can be between 1.0 k and 5.0 kΩ. (For best

temperature coefficient and stability with time, use 1% metal

film resistors).

Vout Vref 1R2

R1

R2 R1Vout

Vref –1



where Vref = 1.23 V

1. Programming Output Voltage (selecting R1 and R2)

Select R1 and R2:

R2 = 9.91 kΩ, choose a 9.88 k metal film resistor.

R2 R1Vout

Vref 11.8 k8.0 V

1.23 V 1

Vout 1.231R2

R1Select R1 = 1.8 kΩ

2. Input Capacitor Selection (Cin)

To prevent large voltage transients from appearing at the input

and for stable operation of the converter, an aluminium or

tantalum electrolytic bypass capacitor is needed between the

input pin +Vin and ground pin Gnd This capacitor should be

located close to the IC using short leads. This capacitor should

have a low ESR (Equivalent Series Resistance) value.

For additional information see input capacitor section in the

“External Components” section of this data sheet.

2. Input Capacitor Selection (Cin)

A 100 µF aluminium electrolytic capacitor located near the

input and ground pin provides sufficient bypassing.

3. Catch Diode Selection (D1)

A.Since the diode maximum peak current exceeds the

regulator maximum load current the catch diode current

rating must be at least 1.2 times greater than the maximum

load current. For a robust design, the diode should have a

current rating equal to the maximum current limit of the

LM2575 to be able to withstand a continuous output short.

B.The reverse voltage rating of the diode should be at least

1.25 times the maximum input voltage.

3. Catch Diode Selection (D1)

A.For this example, a 3.0 A current rating is adequate.

B.Use a 20 V 1N5820 or MBR320 Schottky diode or any

suggested fast recovery diode in the Table 4.

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Procedure (Adjustable Output Version:

LM2575–Adj) (continued)

Procedure Example

4. Inductor Selection (L1)

A.Use the following formula to calculate the inductor Volt x

microsecond [V x µs] constant:

B.Match the calculated E x T value with the corresponding

number on the vertical axis of the Inductor Value Selection

Guide shown in Figure 21. This E x T constant is a measure

of the energy handling capability of an inductor and is

dependent upon the type of core, the core area, the number

of turns, and the duty cycle.

C.Next step is to identify the inductance region intersected by

the E x T value and the maximum load current value on the

horizontal axis shown in Figure 21.

D.From the inductor code, identify the inductor value. Then

select an appropriate inductor from the Table 1 or Table 2.

The inductor chosen must be rated for a switching

frequency of 52 kHz and for a current rating of 1.15 x

IIoad.

The inductor current rating can also be determined by

calculating the inductor peak current

where ton is the “on” time of the power switch and

For additional information about the inductor, see the

inductor section in the “External Components” section of

this data sheet.

ExTVin –V

outVout

Von x106

F[Hz] [V x s]

Ip(max) ILoad(max)Vin –V

outton

ton Vout

Vin x1

fosc

4. Inductor Selection (L1)

A.Calculate E x T [V x µs] constant:

B.E x T = 51 [V x µs]

C.ILoad(max) = 1.0 A

Inductance Region = L220

D.Proper inductor value = 220 µH

Choose the inductor from the Table 1 or Table 2.

ExT(12–8.0

)x8.0

12 x1000

52 51 [V x s]

5. Output Capacitor Selection (Cout)

A.Since the LM2575 is a forward–mode switching regulator

with voltage mode control, its open loop 2–pole–2–zero

frequency characteristic has the dominant pole–pair

determined by the output capacitor and inductor values.

For stable operation, the capacitor must satisfy the

following requirement:

B.Capacitor values between 10 µF and 2000 µF will satisfy

the loop requirements for stable operation. To achieve an

acceptable output ripple voltage and transient response, the

output capacitor may need to be several times larger than

the above formula yields.

C.Due to the fact that the higher voltage electrolytic capacitors

generally have lower ESR (Equivalent Series Resistance)

numbers, the output capacitor’s voltage rating should be at

least 1.5 times greater than the output voltage. For a 5.0 V

regulator, a rating of at least 8V is appropriate, and a 10 V

or 16 V rating is recommended.

Cout 7.785 Vin(max)

Vout xL[µH] [µF]

5. Output Capacitor Selection (Cout)

To achieve an acceptable ripple voltage, select

Cout = 100 µF electrolytic capacitor.

Cout 7.785 12

8.220 53 µF

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INDUCTOR VALUE SELECTION GUIDE

Vin , MAXIMUM INPUT VOLTAGE (V) Vin , MAXIMUM INPUT VOLTAGE (V)

IL, MAXIMUM LOAD CURRENT (A)

0.2

Vin , MAXIMUM INPUT V

LTA

E (V)

Figure 17. LM2575–3.3

IL, MAXIMUM LOAD CURRENT (A)

Figure 18. LM2575–5.0

Figure 19. LM2575–12 Figure 20. LM2575–15

Figure 21. LM2575–Adj

NOTE: This Inductor Value Selection Guide is applicable for continuous mode only.

H1500H1000

L680

L470

L330

L150

H1000

L100

L680

L470

L330

L220

L150

H1500

H1000

H680

H2200

H470

L680 L470

L220

L330

8.0

7.0

6.0

9.0

8.0

7.05.0

0.3

0.30.3

0.4

0.40.4

0.5

0.50.5

0.6

0.60.6

0.7

0.8

0.80.8

0.9

1.0

1.01.0

L220

Vin , MAXIMUM INPUT VOLTAGE (V)

IL, MAXIMUM LOAD CURRENT (A)

0.2

H1500

H1000

H470

H680

H2200

L680

L470

L220

L330

14 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0

0.2

200

ET, VOLTAGE TIME (V s)µ

IL, MAXIMUM LOAD CURRENT (A)

H680

H2200 H1500

H1000

H470

L330

L220

L150

L680

L470

L100

150

125

100

20 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0

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Table 1. Inductor Selection Guide

Inductor

Code Inductor

Value Pulse Eng Renco AIE Tech 39

L100 100 µH PE–92108 RL2444 415–0930 77 308 BV

L150 150 µH PE–53113 RL1954 415–0953 77 358 BV

L220 220 µH PE–52626 RL1953 415–0922 77 408 BV

L330 330 µH PE–52627 RL1952 415–0926 77 458 BV

L470 470 µH PE–53114 RL1951 415–0927 –

L680 680 µH PE–52629 RL1950 415–0928 77 508 BV

H150 150 µH PE–53115 RL2445 415–0936 77 368 BV

H220 220 µH PE–53116 RL2446 430–0636 77 410 BV

H330 330 µH PE–53117 RL2447 430–0635 77 460 BV

H470 470 µH PE–53118 RL1961 430–0634 –

H680 680 µH PE–53119 RL1960 415–0935 77 510 BV

H1000 1000 µH PE–53120 RL1959 415–0934 77 558 BV

H1500 1500 µH PE–53121 RL1958 415–0933 –

H2200 2200 µH PE–53122 RL2448 415–0945 77 610 BV

Table 2. Inductor Selection Guide

Inductance Current Schott Renco Pulse Engineering Coilcraft

(µH) (A) THT SMT THT SMT THT SMT SMT

0.32 67143940 67144310 RL–1284–68–43 RL1500–68 PE–53804 PE–53804–S DO1608–68

0.58 67143990 67144360 RL–5470–6 RL1500–68 PE–53812 PE–53812–S DO3308–683

68 0.99 67144070 67144450 RL–5471–5 RL1500–68 PE–53821 PE–53821–S DO3316–683

1.78 67144140 67144520 RL–5471–5 – PE–53830 PE–53830–S DO5022P–683

0.48 67143980 67144350 RL–5470–5 RL1500–100 PE–53811 PE–53811–S DO3308–104

100 0.82 67144060 67144440 RL–5471–4 RL1500–100 PE–53820 PE–53820–S DO3316–104

1.47 67144130 67144510 RL–5471–4 – PE–53829 PE–53829–S DO5022P–104

0.39 – 67144340 RL–5470–4 RL1500–150 PE–53810 PE–53810–S DO3308–154

150 0.66 67144050 67144430 RL–5471–3 RL1500–150 PE–53819 PE–53819–S DO3316–154

1.20 67144120 67144500 RL–5471–3 – PE–53828 PE–53828–S DO5022P–154

0.32 67143960 67144330 RL–5470–3 RL1500–220 PE–53809 PE–53809–S DO3308–224

220 0.55 67144040 67144420 RL–5471–2 RL1500–220 PE–53818 PE–53818–S DO3316–224

1.00 67144110 67144490 RL–5471–2 – PE–53827 PE–53827–S DO5022P–224

330

0.42 67144030 67144410 RL–5471–1 RL1500–330 PE–53817 PE–53817–S DO3316–334

330 0.80 67144100 67144480 RL–5471–1 – PE–53826 PE–53826–S DO5022P–334

NOTE: Table 1 and Table 2 of this Indicator Selection Guide shows some examples of different manufacturer products suitable for design

with the LM2575.

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Table 3. Example of Several Inductor Manufacturers Phone/Fax Numbers

Pulse Engineering Inc. Phone

Fax + 1–619–674–8100

+ 1–619–674–8262

Pulse Engineering Inc. Europe Phone

Fax + 353 93 24 107

+ 353 93 24 459

Renco Electronics Inc. Phone

Fax + 1–516–645–5828

+ 1–516–586–5562

AIE Magnetics Phone

Fax + 1–813–347–2181

Coilcraft Inc. Phone

Fax + 1–708–322–2645

+ 1–708–639–1469

Coilcraft Inc., Europe Phone

Fax + 44 1236 730 595

+ 44 1236 730 627

Tech 39 Phone

Fax + 33 8425 2626

+ 33 8425 2610

Schott Corp. Phone

Fax + 1–612–475–1173

+ 1–612–475–1786

Table 4. Diode Selection Guide gives an overview about both surface–mount and through–hole diodes for an

effective design. Device listed in bold are available from ON Semiconductor.

Schottky Ultra–Fast Recovery

1.0 A 3.0 A 1.0 A 3.0 A

VRSMT THT SMT THT SMT THT SMT THT

20 V SK12 1N5817

SR102 SK32

MBRD320 1N5820

MBR320

SR302

30 V MBRS130LT3

SK13 1N5818

SR103

11DQ03

SK33

MBRD330 1N5821

MBR330

SR303

31DQ03

MURS120T3 MUR120

11DF1

HER102

MURS320T3

40 V MBRS140T3

SK14

10BQ040

10MQ040

1N5819

SR104

11DQ04

MBRS340T3

MBRD340

30WQ04

SK34

1N5822

MBR340

SR304

31DQ04

10BF10 MURD320 MUR320

30WF10

MUR420

50 V MBRS150

10BQ050 MBR150

SR105

11DQ05

MBRD350

SK35

30WQ05

MBR350

SR305

11DQ05

31DF1

HER302

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EXTERNAL COMPONENTS

Input Capacitor (Cin)

The Input Capacitor Should Have a Low ESR

For stable operation of the switch mode converter a low

ESR (Equivalent Series Resistance) aluminium or solid

tantalum bypass capacitor is needed between the input pin

and the ground pin to prevent large voltage transients from

appearing at the input. It must be located near the regulator

and use short leads. With most electrolytic capacitors, the

capacitance value decreases and the ESR increases with

lower temperatures. For reliable operation in temperatures

below –25°C larger values of the input capacitor may be

needed. Also paralleling a ceramic or solid tantalum

capacitor will increase the regulator stability at cold

temperatures.

RMS Current Rating of C

The important parameter of the input capacitor is the RMS

current rating. Capacitors that are physically large and have

large surface area will typically have higher RMS current

ratings. For a given capacitor value, a higher voltage

electrolytic capacitor will be physically larger than a lower

voltage capacitor, and thus be able to dissipate more heat to

the surrounding air, and therefore will have a higher RMS

current rating. The consequence of operating an electrolytic

capacitor above the RMS current rating is a shortened

operating life. In order to assure maximum capacitor

operating lifetime, the capacitor’ s RMS ripple current rating

should be: Irms > 1.2 x d x ILoad

where d is the duty cycle, for a buck regulator

dton

TVout

Vin

and d ton

T|Vout|

|Vout|V

for a buckboost regulator.

Output Capacitor (Cout)

For low output ripple voltage and good stability, low ESR

output capacitors are recommended. An output capacitor

has two main functions: it filters the output and provides

regulator loop stability. The ESR of the output capacitor and

the peak–to–peak value of the inductor ripple current are the

main factors contributing to the output ripple voltage value.

Standard aluminium electrolytics could be adequate for

some applications but for quality design low ESR types are

recommended.

An aluminium electrolytic capacitor’s ESR value is

related to many factors such as the capacitance value, the

voltage rating, the physical size and the type of construction.

In most cases, the higher voltage electrolytic capacitors have

lower ESR value. Often capacitors with much higher

voltage ratings may be needed to provide low ESR values

that are required for low output ripple voltage.

The Output Capacitor Requires an ESR Value

That Has an Upper and Lower Limit

As mentioned above, a low ESR value is needed for low

output ripple voltage, typically 1% to 2% of the output

voltage. But if the selected capacitor’s ESR is extremely low

(below 0.05 Ω), there is a possibility of an unstable feedback

loop, resulting in oscillation at the output. This situation can

occur when a tantalum capacitor, that can have a very low

ESR, is used as the only output capacitor.

At Low Temperatures, Put in Parallel Aluminium

Electrolytic Capacitors with Tantalum Capacitors

Electrolytic capacitors are not recommended for

temperatures below –25°C. The ESR rises dramatically at

cold temperatures and typically rises 3 times at –25°C and

as much as 10 times at –40°C. Solid tantalum capacitors

have much better ESR spec at cold temperatures and are

recommended for temperatures below –25°C. They can be

also used in parallel with aluminium electrolytics. The value

of the tantalum capacitor should be about 10% or 20% of the

total capacitance. The output capacitor should have at least

50% higher RMS ripple current rating at 52 kHz than the

peak–to–peak inductor ripple current.

Catch Diode

Locate the Catch Diode Close to the LM2575

The LM2575 is a step–down buck converter; it requires a

fast diode to provide a return path for the inductor current

when the switch turns off. This diode must be located close

to the LM2575 using short leads and short printed circuit

traces to avoid EMI problems.

Use a Schottky or a Soft Switching

Ultra–Fast Recovery Diode

Since the rectifier diodes are very significant source of

losses within switching power supplies, choosing the

rectifier that best fits into the converter design is an

important process. Schottky diodes provide the best

performance because of their fast switching speed and low

forward voltage drop.

They provide the best efficiency especially in low output

voltage applications (5.0 V and lower). Another choice

could be Fast–Recovery, or Ultra–Fast Recovery diodes. It

has to be noted, that some types of these diodes with an

abrupt turnoff characteristic may cause instability or EMI

troubles.

A fast–recovery diode with soft recovery characteristics

can better fulfill a quality, low noise design requirements.

Table 4 provides a list of suitable diodes for the LM2575

regulator. Standard 50/60 Hz rectifier diodes such as the

1N4001 series or 1N5400 series are NOT suitable.

Inductor

The magnetic components are the cornerstone of all

switching power supply designs. The style of the core and

the winding technique used in the magnetic component’s

design has a great influence on the reliability of the overall

power supply.

Using an improper or poorly designed inductor can cause

high voltage spikes generated by the rate of transitions in

current within the switching power supply, and the

possibility of core saturation can arise during an abnormal

operational mode. Voltage spikes can cause the

semiconductors to enter avalanche breakdown and the part

can instantly fail if enough energy is applied. It can also

cause significant RFI (Radio Frequency Interference) and

EMI (Electro–Magnetic Interference) problems.

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Continuous and Discontinuous Mode of Operation

The LM2575 step–down converter can operate in both the

continuous and the discontinuous modes of operation. The

regulator works in the continuous mode when loads are

relatively heavy, the current flows through the inductor

continuously and never falls to zero. Under light load

conditions, the circuit will be forced to the discontinuous

mode when inductor current falls to zero for certain period

of time (see Figure 22 and Figure 23). Each mode has

distinctively different operating characteristics, which can

affect the regulator performance and requirements. In many

cases the preferred mode of operation is the continuous

mode. It o ffers greater output power, lower peak currents in

the switch, inductor and diode, and can have a lower output

ripple voltage. On the other hand it does require larger

inductor values to keep the inductor current flowing

continuously, especially at low output load currents and/or

high input voltages.

To simplify the inductor selection process, an inductor

selection guide for the LM2575 regulator was added to this

data sheet (Figures 17 through 21). This guide assumes that

the regulator is operating in the continuous mode, and

selects an inductor that will allow a peak–to–peak inductor

ripple current to be a certain percentage of the maximum

design load current. This percentage is allowed to change as

different design load currents are selected. For light loads

(less than approximately 200 mA) it may be desirable to

operate the regulator in the discontinuous mode, because the

inductor value and size can be kept relatively low.

Consequently, the percentage of inductor peak–to–peak

current increases. This discontinuous mode of operation is

perfectly acceptable for this type of switching converter.

Any buck regulator will be forced to enter discontinuous

mode if the load current is light enough.

Figure 22. Continuous Mode Switching

Current Waveforms

POWER SWITCH

1.0

CURRENT (A)

HORIZONTAL TIME BASE: 5.0 µs/DIV

1.0

INDUCTOR

CURRENT (A)

Selecting the Right Inductor Style

Some important considerations when selecting a core type

are core material, cost, the output power of the power supply,

the physical volume the inductor must fit within, and the

amount of EMI (Electro–Magnetic Interference) shielding

that the core must provide. The inductor selection guide

covers d i fferent styles of inductors, such as pot core, E–core,

toroid and bobbin core, as well as different core materials

such as ferrites and powdered iron from different

manufacturers.

For hi g h quality design regulators the toroid core seems to

be the best choice. Since the magnetic flux is completely

contained within the core, it generates less EMI, reducing

noise problems in sensitive circuits. The least expensive is

the bobbin core type, which consists of wire wound on a

ferrite rod core. This type of inductor generates more EMI

due to the fact that its core is open, and the magnetic flux is

not completely contained within the core.

When multiple switching regulators are located on the

same printed circuit board, open core magnetics can cause

interference between two or more of the regulator circuits,

especially a t high currents due to mutual coupling. A toroid,

pot core or E–core (closed magnetic structure) should be

used in such applications.

Do Not Operate an Inductor Beyond its

Maximum Rated Current

Exceeding an inductor’s maximum current rating may

cause the inductor to overheat because of the copper wire

losses, o r the core may saturate. Core saturation occurs when

the flux density is too high and consequently the cross

sectional area of the core can no longer support additional

lines of magnetic flux.

This causes the permeability of the core to drop, the

inductance value decreases rapidly and the inductor begins

to look mainly resistive. It has only the dc resistance of the

winding. This can cause the switch current to rise very

rapidly and force the LM2575 internal switch into

cycle–by–cycle current limit, thus reducing the dc output

load current. This can also result in overheating of the

inductor and/or the LM2575. Different inductor types have

different saturation characteristics, and this should be kept

in mind when selecting an inductor.

Figure 23. Discontinuous Mode Switching

Current Waveforms

0.1

HORIZONTAL TIME BASE: 5.0 µs/DIV

POWER SWITCH

CURRENT (A)

INDUCTOR

CURRENT (A)

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GENERAL RECOMMENDATIONS

Output Voltage Ripple and Transients

Source of the Output Ripple

Since the LM2575 is a switch mode power supply

regulator, its output voltage, if left unfiltered, will contain a

sawtooth ripple voltage at the switching frequency. The

output ripple voltage value ranges from 0.5% to 3% of the

output voltage. It is caused mainly by the inductor sawtooth

ripple current multiplied by the ESR of the output capacitor.

Short Voltage Spikes and How to Reduce Them

The regulator output voltage may also contain short

voltage spikes at the peaks of the sawtooth waveform (see

Figure 24). These voltage spikes are present because of the

fast switching action of the output switch, and the parasitic

inductance of the output filter capacitor. There are some

other important factors such as wiring inductance, stray

capacitance, a s well as the scope probe used to evaluate these

transients, all these contribute to the amplitude of these

spikes. To minimize these voltage spikes, low inductance

capacitors should be used, and their lead lengths must be

kept short. The importance of quality printed circuit board

layout design should also be highlighted.

Figure 24. Output Ripple Voltage Waveforms

HORIZONTAL TIME BASE: 10 µs/DIV

UNFILTERED

OUTPUT

VOLTAGE

VERTICAL

RESOLUTION:

20 mV/DIV

FILTERED

OUTPUT

VOLTAGE

Voltage spikes caused by switching action of the output

switch and the parasitic inductance of the output capacitor

Minimizing the Output Ripple

In order to minimize the output ripple voltage it is possible

to enlarge the inductance value of the inductor L1 and/or to

use a l a rger value output capacitor. There is also another way

to smooth the output by means of an additional LC filter

(20 µH, 100 µF), that can be added to the output (see

Figure 33) to further reduce the amount of output ripple and

transients. With such a filter it is possible to reduce the

output ripple voltage transients 10 times or more. Figure 24

shows the difference between filtered and unfiltered output

waveforms of the regulator shown in Figure 33.

The upper waveform is from the normal unfiltered output

of the converter , while the lower waveform shows the output

ripple voltage filtered by an additional LC filter.

Heatsinking and Thermal Considerations

The Through–Hole Package TO–220

The LM2575 is available in two packages, a 5–pin

TO–220(T, TV) and a 5–pin surface mount D2PAK(D2T).

There are many applications that require no heatsink to keep

the LM2575 junction temperature within the allowed

operating range. The TO–220 package can be used without

a heatsink for ambient temperatures up to approximately

50°C (depending on the output voltage and load current).

Higher ambient temperatures require some heatsinking,

either to the printed circuit (PC) board or an external

heatsink.

The Surface Mount Package D

PAK

and its

Heatsinking

The other type of package, the surface mount D2PAK, is

designed to be soldered to the copper on the PC board. The

copper and the board are the heatsink for this package and

the other heat producing components, such as the catch

diode and inductor. The PC board copper area that the

package i s soldered to should be at least 0.4 in2 (or 100 mm2)

and ideally should have 2 or more square inches (1300 mm2)

of 0.0028 inch copper. Additional increasing of copper area

beyond approximately 3.0 in2 (2000 mm2) will not improve

heat dissipation significantly. If further thermal

improvements are needed, double sided or multilayer PC

boards with large copper areas should be considered.

Thermal Analysis and Design

The following procedure must be performed to determine

whether or not a heatsink will be required. First determine:

1. PD(max) maximum regulator power dissipation in

the application.

2. TA(max) maximum ambient temperature in the

application.

3. TJ(max) maximum allowed junction temperature

(125°C for the LM2575). For a conservative

design, the maximum junction temperature

should not exceed 110°C to assure safe

operation. For every additional 10°C

temperature rise that the junction must

withstand, the estimated operating lifetime

of the component is halved.

4. RθJC package thermal resistance junction–case.

5. RθJA package thermal resistance junction–ambient.

(Refer to Absolute Maximum Ratings in this data sheet or

RθJC and RθJA values).

The following formula is to calculate the total power

dissipated by the LM2575:

PD = (Vin x IQ) + d x ILoad x Vsat

where d is the duty cycle and for buck converter

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dton

TVO

Vin,

IQ(quiescent current) and Vsat can be found in the

LM2575 data sheet,

Vin is minimum input voltage applied,

VOis the regulator output voltage,

ILoad is the load current.

The dynamic switching losses during turn–on and

turn–off can be neglected if proper type catch diode is used.

Packages Not on a Heatsink (Free–Standing)

For a free–standing application when no heatsink is used,

the junction temperature can be determined by the following

expression:

TJ = (RθJA) (PD) + TA

where (RθJA)(PD) represents the junction temperature rise

caused by the dissipated power and TA is the maximum

ambient temperature.

Packages on a Heatsink

If the actual operating junction temperature is greater than

the selected safe operating junction temperature determined

in step 3, than a heatsink is required. The junction

temperature will be calculated as follows:

TJ = PD (RθJA + RθCS + RθSA) + TA

where RθJC is the thermal resistance junction–case,

RθCS is the thermal resistance case–heatsink,

RθSA is the thermal resistance heatsink–ambient.

If the actual operating temperature is greater than the

selected safe operating junction temperature, then a larger

heatsink is required.

Some Aspects That can Influence Thermal Design

It should be noted that the package thermal resistance and

the junction temperature rise numbers are all approximate,

and there are many factors that will affect these numbers,

such as PC board size, shape, thickness, physical position,

location, board temperature, as well as whether the

surrounding air is moving or still.

Other factors are trace width, total printed circuit copper

area, copper thickness, single– or double–sided, multilayer

board, the amount of solder on the board or even color of the

traces.

The s ize, quantity and spacing of other components on

the board can also influence its effectiveness to dissipate

the heat.

Figure 25. Inverting Buck–Boost Regulator Using the

LM2575–12 Develops –12 V @ 0.35 A

1N5819

100 µH

Output

Feedback

Unregulated

DC Input

12 V to 25 V

Cin

100 µF

/50 V

53ON

/OFFGnd

+Vin

Regulated

Output

-12 V @ 0.35 A

Cout

1800 µF

/16 V

LM2575–12

ADDITIONAL APPLICATIONS

Inverting Regulator

An inverting buck–boost regulator using the LM2575–12

is shown in Figure 25. This circuit converts a positive input

voltage to a negative output voltage with a common ground

by bootstrapping the regulators ground to the negative

output voltage. By grounding the feedback pin, the regulator

senses the inverted output voltage and regulates it.

In this example the LM2575–12 is used to generate a

–12 V output. The maximum input voltage in this case

cannot exceed +28 V because the maximum voltage

appearing across the regulator is the absolute sum of the

input and output voltages and this must be limited to a

maximum of 40 V.

This circuit configuration is able to deliver approximately

0.35 A t o the output when the input voltage is 12 V or higher .

At lighter loads the minimum input voltage required drops

to approximately 4.7 V, because the buck–boost regulator

topology can produce an output voltage that, in its absolute

value, is either greater or less than the input voltage.

Since the switch currents in this buck–boost configuration

are higher than in the standard buck converter topology, the

available output current is lower.

This type of buck–boost inverting regulator can also

require a larger amount of startup input current, even for

light loads. This may overload an input power source with

a current limit less than 1.5 A.

Such an amount of input startup current is needed for at

least 2.0 ms or more. The actual time depends on the output

voltage and size of the output capacitor.

Because o f the relatively high startup currents required by

this inverting regulator topology, the use of a delayed startup

or an undervoltage lockout circuit is recommended.

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Using a delayed startup arrangement, the input capacitor

can charge up to a higher voltage before the switch–mode

regulator begins to operate.

The high input current needed for startup is now partially

supplied by the input capacitor Cin.

Design Recommendations:

The inverting regulator operates in a different manner

than the buck converter and so a different design procedure

has to be used to select the inductor L1 or the output

capacitor Cout.

The output capacitor values must be larger than is

normally required for buck converter designs. Low input

voltages or high output currents require a large value output

capacitor (in the range of thousands of µF).

The recommended range of inductor values for the

inverting converter design is between 68 µH and 220 µH. To

select an inductor with an appropriate current rating, the

inductor peak current has to be calculated.

The following formula is used to obtain the peak inductor

current:

where ton |VO|

Vin |VO|x1

fosc, and fosc 52 kHz.

Ipeak ILoad (Vin |VO|)

Vin Vin xt

2L1

Under normal continuous inductor current operating

conditions, the worst case occurs when Vin is minimal.

Note that the voltage appearing across the regulator is the

absolute sum of the input and output voltage, and must not

exceed 40 V.

Figure 26. Inverting Buck–Boost

Regulator with Delayed Startup

1N5819

100 µH

Output

Feedback

Unregulated

DC Input

12 V to 25 V

Cin

100 µF

/50 V

35ON

/OFF Gnd

+Vin

Regulated

Output

-12 V @ 0.35 A

Cout

1800 µF

/16 V

LM2575–12

0.1 µF

47 k R2

47 k

It has been already mentioned above, that in some

situations, the delayed startup or the undervoltage lockout

features could be very useful. A delayed startup circuit

applied to a buck–boost converter is shown in Figure 26.

Figure 32 in the “Undervoltage Lockout” section describes

an undervoltage lockout feature for the same converter

topology.

Figure 27. Inverting Buck–Boost Regulator Shut Dow

Circuit Using an Optocoupler

LM2575–XX

35 GndON/OFF

+Vin

47 k

Cin

100 µF

NOTE: This picture does not show the complete circuit.

47 k

470

Shutdown

Input

MOC8101

-Vout

Off

5.0 V

+Vin

With the inverting configuration, the use of the ON/OFF

pin requires some level shifting techniques. This is caused

by the fact, that the ground pin of the converter IC is no

longer at ground. Now, the ON/OFF pin threshold voltage

(1.4 V approximately) has to be related to the negative

output voltage level. There are many different possible shut

down methods, two of them are shown in Figures 27 and 28.

Figure 28. Inverting Buck–Boost Regulator Shut Dow

Circuit Using a PNP Transistor

NOTE: This picture does not show the complete circuit.

5.6 k

2N3906

LM2575–XX

35 GndON/OFF

12 k -Vout

+Vin

Shutdown

Input

Off

+Vin

Cin

100 µF

Negative Boost Regulator

This example is a variation of the buck–boost topology

and is called a negative boost regulator. This regulator

experiences relatively high switch current, especially at low

input voltages. The internal switch current limiting results in

lower output load current capability.

The circuit in Figure 29 shows the negative boost

configuration. The input voltage in this application ranges

from –5.0 V to –12 V and provides a regulated –12 V output.

LM2575

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If the input voltage is greater than –12 V, the output will rise

above –12 V accordingly, but will not damage the regulator.

Figure 29. Negative Boost Regulator

1N5817

150 µH

Output

Feedback

Regulated

Output

Vout = -12 V

Load Current from

200 mA for Vin = -5.2 V

to 500 mA for Vin = -7.0 V

Unregulated

DC Input

-Vin = -5.0 V to -12 V

Cout

1000 µF

/16 V

Cin

100 µF

/50 V

LM2575–12

53 ON/OFFGnd

+Vin

Design Recommendations:

The same design rules as for the previous inverting

buck–boost converter can be applied. The output capacitor

Cout must be chosen larger than would be required for a

standard buck converter. Low input voltages or high output

currents require a large value output capacitor (in the range

of thousands of µF). The recommended range of inductor

values for the negative boost regulator is the same as for

inverting converter design.

Another important point is that these negative boost

converters cannot provide current limiting load protection in

the event of a short in the output so some other means, such

as a fuse, may be necessary to provide the load protection.

Delayed Startup

There are some applications, like the inverting regulator

already mentioned above, which require a higher amount of

startup current. In such cases, if the input power source is

limited, this delayed startup feature becomes very useful.

To provide a time delay between the time the input voltage

is applied and the time when the output voltage comes up,

the circuit in Figure 30 can be used. As the input voltage is

applied, the capacitor C1 charges up, and the voltage across

the resistor R2 falls down. When the voltage on the ON/OFF

pin falls below the threshold value 1.4 V, the regulator starts

up. Resistor R1 is included to limit the maximum voltage

applied to t h e O N/OFF pin, reduces the power supply noise

sensitivity, and also limits the capacitor C1 discharge

current, but its use is not mandatory.

When a high 50 Hz or 60 Hz (100 Hz or 120 Hz

respectively) ripple voltage exists, a long delay time can

cause some problems by coupling the ripple into the

ON/OFF pin, the regulator could be switched periodically

on and off with the line (or double) frequency.

Figure 30. Delayed Startup Circuitry

47 k

LM2575–XX

35 GndON/OFF

47 k

+Vin

0.1 µF

Cin

100 µF

NOTE: This picture does not show the complete circuit.

Undervoltage Lockout

Some applications require the regulator to remain of f until

the input voltage reaches a certain threshold level. Figure 31

shows an undervoltage lockout circuit applied to a buck

regulator. A version of this circuit for buck–boost converter

is shown in Figure 32. Resistor R3 pulls the ON/OFF pin

high and keeps the regulator off until the input voltage

reaches a predetermined threshold level, which is

determined by the following expression:

Vth VZ1 1R2

R1VBE (Q1)

Figure 31. Undervoltage Lockout Circuit for

Buck Converter

10 k

1N5242B

10 k

2N3904

47 k

Vth ≈ 13 V

Cin

100 µF

LM2575–5.0

35 GndON/OFF

+Vin

NOTE: This picture does not show the complete circuit.

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Figure 32. Undervoltage Lockout Circuit for

Buck–Boost Converter

15 k

1N5242B

15 k

2N3904

68 k

Vth ≈ 13 V

Cin

100 µF

LM2575–5.0

35 GndON/OFF

+Vin

Vout = -5.0 V

NOTE: This picture does not show the complete circuit.

Adjustable Output, Low–Ripple Power Supply

A 1.0 A output current capability power supply that

features an adjustable output voltage is shown in Figure 33.

This regulator delivers 1.0 A into 1.2 V to 35 V output.

The input voltage ranges from roughly 8.0 V to 40 V. In order

to achieve a 10 or more times reduction of output ripple, an

additional L–C filter is included in this circuit.

Figure 33. Adjustable Power Supply with Low Ripple Voltage

1N5819

150 µH

Output

Feedback

50 k

1.1 k

20 µHRegulated

Output Voltage

1.2 V to 35 V @1.0 A

Optional Output

Ripple Filter

Unregulated

DC Input

Cout

2200 µF

100 µF

Cin

100 µF

/50 V

LM2575–Adj

53ON

/OFFGnd

+Vin

R , THERMAL RESISTANCE

JAθ

JUNCTIONTOAIR ( C/W)°

1.0

1.5

2.0

2.5

3.0

3.5

010203025155.0

L, LENGTH OF COPPER (mm)

Minimum

Size Pad

2.0 oz. Copper

ÎÎÎÎ

Free Air

Mounted

Vertically

PD, MAXIMUM POWER DISSIPATION (W)

Figure 34. D2PAK Thermal Resistance and Maximum

Power Dissipation versus P.C.B. Copper Length

PD(max) for TA = 50°C

RθJA

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THE LM2575–5.0 STEP–DOWN VOLTAGE REGULATOR WITH 5.0 V @ 1.0 A OUTPUT POWER

CAPABILITY. TYPICAL APPLICATION WITH THROUGH–HOLE PC BOARD LAYOUT

DC-DC Converter

Figure 35. Schematic Diagram of the LM2575–5.0 Step–Down Converter

Figure 36. Printed Circuit Board

Component Side Figure 37. Printed Circuit Board

Copper Side

1N5819

330 µH

Output

Feedback

Unregulated

DC Input

+Vin = +7.0 V to +40 V

Cout

330 µF

/16 V

100 µF

/50 V

LM2575–5.0

53ON

/OFFGnd

+Vin

Regulated Output

+Vout1 = 5.0 V @ 1.0 A

Gndin Gndout

C1 – 100 µF, 50 V, Aluminium Electrolytic

C2 – 330 µF, 16 V, Aluminium Electrolytic

D1 – 1.0 A, 40 V, Schottky Rectifier, 1N5819

L1 – 330 µH, Tech 39: 77 458 BV, Toroid Core, Through–Hole, Pin 3 = Start, Pin 7 = Finish

NOTE: Not to scale. NOTE: Not to scale.

+Vout1

+Vin

Gndin Gndout

U1 LM2575

LM2575

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THE LM2575–ADJ STEP–DOWN VOLTAGE REGULATOR WITH 8.0 V @ 1.0 A OUTPUT POWER

CAPABILITY. TYPICAL APPLICATION WITH THROUGH–HOLE PC BOARD LAYOUT

C1 – 100 µF, 50 V, Aluminium Electrolytic

C2 – 330 µF, 16 V, Aluminium Electrolytic

C3 – 100 µF, 16 V, Aluminium Electrolytic

D1 – 1.0 A, 40 V, Schottky Rectifier, 1N5819

L1 – 330 µH, Tech 39: 77 458 BV, Toroid Core, Through–Hole, Pin 3 = Start, Pin 7 = Finish

L2 – 25 µH, TDK: SFT52501, Toroid Core, Through–Hole

R1 – 1.8 k

R2 – 10 k

Figure 38. Schematic Diagram of the 8.0 V @ 1.0 V Step–Down Converter Using the LM2575–Adj

(An additional LC filter is included to achieve low output ripple voltage)

Figure 39. PC Board Component Side Figure 40. PC Board Copper Side

Vref = 1.23 V

R1 is between 1.0 k and 5.0 k

1N5819

330 µH

Output

10 k

1.8 k

25 µHRegulated

Output Filtered

Vout2 = 8.0 V @1.0 A

Unregulated

DC Input

330 µF

/16 V

100 µF

/16 V

100 µF

/50 V

LM2575–Adj

53ON

/OFFGnd

+Vin

+Vin = +10 V to + 40 V

4 Feedback

Regulated

Output Unfiltered

Vout1 = 8.0 V @1.0 A

Vout Vref 1R2

R1

+Vout1

+Vin

Gndin

D1 J1

U1 LM2575

+Vout2

R2 R1

Gndout

NOTE: Not to scale. NOTE: Not to scale.

References

•National Semiconductor LM2575 Data Sheet and Application Note

•National Semiconductor LM2595 Data Sheet and Application Note

•Marty Brown “Practical Switching Power Supply Design”, Academic Press, Inc., San Diego 1990

•Ray Ridley “High Frequency Magnetics Design”, Ridley Engineering, Inc. 1995

LM2575

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ORDERING INFORMATION

Device Nominal

Output Voltage Operating

Temperature Range Package Shipping

LM2575TV–ADJ TO–220 (Vertical Mount)

LM2575T–ADJ

123Vto37V

TO–220 (Straight Lead)

LM2575D2T–ADJ 1.23 V to 37 V D2PAK (Surface Mount)

LM2575D2T–ADJR4 D2PAK (Surface Mount)

LM2575TV–3.3 TO–220 (Vertical Mount)

LM2575T–3.3

33V

TO–220 (Straight Lead) 50 Units/Rail

LM2575D2T–3.3 3.3 V D2PAK (Surface Mount)

LM2575D2T–3.3R4 D2PAK (Surface Mount)

LM2575TV–5 TO–220 (Vertical Mount)

LM2575T–5

50V

T40°to +125°C

TO–220 (Straight Lead)

LM2575D2T–5 5.0 V TJ = –40° to +125°CD2PAK (Surface Mount)

LM2575D2T–5R4 D2PAK (Surface Mount) 800 Tape & Reel

LM2575TV–12 TO–220 (Vertical Mount)

LM2575T–12

12 V

TO–220 (Straight Lead)

LM2575D2T–12 12 V D2PAK (Surface Mount)

LM2575D2T–12R4 D2PAK (Surface Mount)

50 Units/Rail

LM2575TV–15 TO–220 (Vertical Mount) 50 Units/Rail

LM2575T–15

15 V

TO–220 (Straight Lead)

LM2575D2T–15 15 V D2PAK (Surface Mount)

LM2575D2T–15R4 D2PAK (Surface Mount)

xxx = 3.3, 5.0, 12, 15, or ADJ

A = Assembly Location

WL = Wafer Lot

Y = Year

WW = Work Week

TO–220

TV SUFFIX

CASE 314B

MARKING DIAGRAMS

TO–220

T SUFFIX

CASE 314D

TO–220

T SUFFIX

CASE 314D

D2PAK

D2T SUFFIX

CASE 936A

2575T–xxx

AWLYWW

2575T–xxx

AWLYWW

2575–xxx

AWLYWW

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PACKAGE DIMENSIONS

TO–220

TV SUFFIX

CASE 314B–05

ISSUE J

–P–

0.10 (0.254) P M

5X J

0.24 (0.610) T

OPTIONAL

CHAMFER

SLW

–T– SEATING

PLANE

NOTES:

1. DIMENSIONING AND TOLERANCING PER ANSI

Y14.5M, 1982.

2. CONTROLLING DIMENSION: INCH.

3. DIMENSION D DOES NOT INCLUDE

INTERCONNECT BAR (DAMBAR) PROTRUSION.

DIMENSION D INCLUDING PROTRUSION SHALL

NOT EXCEED 0.043 (1.092) MAXIMUM.

DIM MIN MAX MIN MAX

MILLIMETERSINCHES

A0.572 0.613 14.529 15.570

B0.390 0.415 9.906 10.541

C0.170 0.180 4.318 4.572

D0.025 0.038 0.635 0.965

E0.048 0.055 1.219 1.397

F0.850 0.935 21.590 23.749

G0.067 BSC 1.702 BSC

H0.166 BSC 4.216 BSC

J0.015 0.025 0.381 0.635

K0.900 1.100 22.860 27.940

L0.320 0.365 8.128 9.271

N0.320 BSC 8.128 BSC

Q0.140 0.153 3.556 3.886

S--- 0.620 --- 15.748

U0.468 0.505 11.888 12.827

V--- 0.735 --- 18.669

W0.090 0.110 2.286 2.794

5X D

TO–220

T SUFFIX

CASE 314D–04

ISSUE E

–Q–

12345

5 PL

0.356 (0.014) T

SEATING

PLANE

–T–

DIM MIN MAX MIN MAX

MILLIMETERSINCHES

A0.572 0.613 14.529 15.570

B0.390 0.415 9.906 10.541

C0.170 0.180 4.318 4.572

D0.025 0.038 0.635 0.965

E0.048 0.055 1.219 1.397

G0.067 BSC 1.702 BSC

H0.087 0.112 2.210 2.845

J0.015 0.025 0.381 0.635

K0.990 1.045 25.146 26.543

L0.320 0.365 8.128 9.271

Q0.140 0.153 3.556 3.886

U0.105 0.117 2.667 2.972

NOTES:

1. DIMENSIONING AND TOLERANCING PER ANSI

Y14.5M, 1982.

2. CONTROLLING DIMENSION: INCH.

3. DIMENSION D DOES NOT INCLUDE

INTERCONNECT BAR (DAMBAR) PROTRUSION.

DIMENSION D INCLUDING PROTRUSION SHALL

NOT EXCEED 10.92 (0.043) MAXIMUM.

LM2575

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PACKAGE DIMENSIONS

D2PAK

D2T SUFFIX

CASE 936A–02

ISSUE B

5 REF

123

TERMINAL 6

NOTES:

1. DIMENSIONING AND TOLERANCING PER ANSI

Y14.5M, 1982.

2. CONTROLLING DIMENSION: INCH.

3. TAB CONTOUR OPTIONAL WITHIN DIMENSIONS

A AND K.

4. DIMENSIONS U AND V ESTABLISH A MINIMUM

MOUNTING SURFACE FOR TERMINAL 6.

5. DIMENSIONS A AND B DO NOT INCLUDE MOLD

FLASH OR GATE PROTRUSIONS. MOLD FLASH

AND GATE PROTRUSIONS NOT TO EXCEED

0.025 (0.635) MAXIMUM.

DIM

MIN MAX MIN MAX

MILLIMETERS

0.386 0.403 9.804 10.236

INCHES

B0.356 0.368 9.042 9.347

C0.170 0.180 4.318 4.572

D0.026 0.036 0.660 0.914

E0.045 0.055 1.143 1.397

G0.067 BSC 1.702 BSC

H0.539 0.579 13.691 14.707

K0.050 REF 1.270 REF

L0.000 0.010 0.000 0.254

M0.088 0.102 2.235 2.591

N0.018 0.026 0.457 0.660

P0.058 0.078 1.473 1.981

R5 REF

S0.116 REF 2.946 REF

U0.200 MIN 5.080 MIN

V0.250 MIN 6.350 MIN



0.010 (0.254) T

–T–

OPTIONAL

CHAMFER

LM2575

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Notes

LM2575

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ON Semiconductor and are trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes

without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular

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including without limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or

specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals” must be

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For additional information, please contact your local

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LM2575/D

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