REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
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may result from its use. No license is granted by implication or otherwise
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One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700 www.analog.com
Fax: 781/326-8703 © 2003 Analog Devices, Inc. All rights reserved.
AD8628
Zero-Drift, Single-Supply Rail-to-Rail
Input/Output Operational Amplifier
PIN CONFIGURATIONS
5-Lead SOT-23
(RJ Suffix)
1
2
3
5
4–IN
+IN
V+
OUT
AD8628
V–
8-Lead SOIC
(R Suffix)
1
2
3
4
8
7
6
5
AD8628
–IN
V–
+IN
V+
OUT
NC
NC
NC
NC = NO CONNECT
FEATURES
Lowest Auto-Zero Amplifier Noise
Low Offset Voltage: 1 ␮V
Input Offset Drift: 0.02 ␮V/ⴗC
Rail-to-Rail Input and Output Swing
5 V Single-Supply Operation
High Gain, CMRR, and PSRR: 120 dB
Very Low Input Bias Current: 100 pA
Low Supply Current: 1.0 mA
Overload Recovery Time: 10 ␮s
No External Components Required
APPLICATIONS
Automotive Sensors
Pressure and Position Sensors
Strain Gage Amplifiers
Medical Instrumentation
Thermocouple Amplifiers
Precision Current Sensing
Photodiode Amplifier
GENERAL DESCRIPTION
This new breed of amplifier has ultralow offset, drift, and bias
current. The AD8628 is a wide bandwidth auto-zero amplifier
featuring rail-to-rail input and output swings and low noise.
Operation is fully specified from 2.7 V to 5 V single supply
(±1.35 V to ±2.5 V dual supply).
The AD8628 family provides benefits previously found only in
expensive auto-zeroing or chopper-stabilized amplifiers. Using
Analog Devices’ new topology, these zero-drift amplifiers com-
bine low cost with high accuracy and low noise. (No external
capacitors are required.) In addition, the AD8628 greatly reduces
the digital switching noise found in most chopper-stabilized
amplifiers.
With an offset voltage of only 1 mV, drift less than 0.005 mV/∞C,
and noise of only 0.5 mV p-p (0 Hz to 10 Hz), the AD8628 is
perfectly suited for applications where error sources cannot be
tolerated. Position and pressure sensors, medical equipment,
and strain gage amplifiers benefit greatly from nearly zero drift
over their operating temperature range. Many systems can take
advantage of the rail-to-rail input and output swings provided
by the AD8628 family to reduce input biasing complexity and
maximize SNR.
The AD8628 family is specified for the extended industrial tempera-
ture range (–40∞C to +125∞C). The AD8628 amplifier is available
in the tiny SOT-23 and the popular 8-lead narrow SOIC plastic
packages. The SOT-23 package devices are available only in
tape and reel.
REV. A–2–
AD8628–SPECIFICATIONS
ELECTRICAL CHARACTERISTICS
Parameter Symbol Conditions Min Typ Max Unit
INPUT CHARACTERISTICS
Offset Voltage V
OS
15 mV
–40∞C £ T
A
£ +125∞C10mV
Input Bias Current I
B
30 100 pA
–40∞C £ T
A
£ +125∞C1.5 nA
Input Offset Current I
OS
50 200 pA
–40∞C £ T
A
£ +125∞C250 pA
Input Voltage Range 05V
Common-Mode Rejection Ratio CMRR V
CM
= 0 V to 5 V 120 140 dB
–40∞C £ T
A
£ +125∞C115 130 dB
Large Signal Voltage Gain*A
VO
R
L
= 10 kW, V
O
= 0.3 V to 4.7 V 125 145 dB
–40∞C £ T
A
£ +125∞C120 135 dB
Offset Voltage Drift DV
OS
/DT –40∞C £ T
A
£ +125∞C0.002 0.02 mV/∞C
OUTPUT CHARACTERISTICS
Output Voltage High V
OH
R
L
= 100 kW to Ground 4.99 4.996 V
–40∞C £ T
A
£ +125∞C4.99 4.995 V
R
L
= 10 kW to Ground 4.95 4.98 V
–40∞C £ T
A
£ +125∞C4.95 4.97 V
Output Voltage Low V
OL
R
L
= 100 kW to V+ 1 5 mV
–40∞C £ T
A
£ +125∞C25mV
R
L
= 10 kW to V+ 10 20 mV
–40∞C £ T
A
£ +125∞C1520mV
Short-Circuit Limit I
SC
± 25 ± 50 mA
–40∞C £ T
A
£ +125∞C± 40 mA
Output Current I
O
± 30 mA
–40∞C £ T
A
£ +125∞C± 15 mA
POWER SUPPLY
Power Supply Rejection Ratio PSRR V
S
= 2.7 V to 5.5 V
–40∞C £ T
A
£ +125∞C115 130 dB
Supply Current/Amplifier I
SY
V
O
= 0 V 0.85 1.1 mA
–40∞C £ T
A
£ +125∞C1.0 1.2 mA
INPUT CAPACITANCE
Differential C
IN
1.5 pF
Common Mode 10 pF
DYNAMIC PERFORMANCE
Slew Rate SR R
L
= 10 kW1.0 V/ms
Overload Recovery Time 0.05 ms
Gain Bandwidth Product GBP 2.5 MHz
NOISE PERFORMANCE
Voltage Noise e
n
p-p 0.1 Hz to 10 Hz 0.5 mV p-p
e
n
p-p 0.1 Hz to 1.0 Hz 0.16 mV p-p
Voltage Noise Density e
n
f = 1 kHz 22 nV/÷Hz
Current Noise Density i
n
f = 10 Hz 5 fA/÷Hz
*Gain testing is highly dependent upon test bandwidth.
Specifications subject to change without notice.
(VS = 5.0 V, VCM = 2.5 V, TA = 25ⴗC, unless otherwise noted.)
REV. A
AD8628
–3–
ELECTRICAL CHARACTERISTICS
Parameter Symbol Conditions Min Typ Max Unit
INPUT CHARACTERISTICS
Offset Voltage V
OS
15 mV
–40∞C £ T
A
£ +125∞C10mV
Input Bias Current I
B
30 100 pA
–40∞C £ T
A
£ +125∞C1.0 1.5 nA
Input Offset Current I
OS
50 200 pA
–40∞C £ T
A
£ +125∞C250 pA
Input Voltage Range 05V
Common-Mode Rejection Ratio CMRR V
CM
= 0 V to 2.7 V 115 130 dB
–40∞C £ T
A
£ +125∞C110 120 dB
Large Signal Voltage Gain A
VO
R
L
= 10 kW , V
O
= 0.3 V to 2.4 V 110 140 dB
–40∞C £ T
A
£ +125∞C105 130 dB
Offset Voltage Drift DV
OS
/DT–40∞C £ T
A
£ +125∞C0.002 0.02 mV/∞C
OUTPUT CHARACTERISTICS
Output Voltage High V
OH
R
L
= 100 kW to Ground 2.68 2.695 V
–40∞C £ T
A
£ +125∞C2.68 2.695 V
R
L
= 10 kW to Ground 2.67 2.68 V
–40∞C £ T
A
£ +125∞C2.67 2.675 V
Output Voltage Low V
OL
R
L
= 100 kW to V+ 1 5 mV
–40∞C £ T
A
£ +125∞C25mV
R
L
= 10 kW to V+ 10 20 mV
–40∞C £ T
A
£ +125∞C1520mV
Short-Circuit Limit I
SC
±10 ±15 mA
–40∞C £ T
A
£ +125∞C±10 mA
Output Current I
O
±10 mA
–40∞C £ T
A
£ +125∞C±5mA
POWER SUPPLY
Power Supply Rejection Ratio PSRR V
S
= 2.7 V to 5.5 V
–40∞C £ T
A
£ +125∞C115 130 dB
Supply Current/Amplifier I
SY
V
O
= 0 V 0.75 1.0 mA
–40∞C £ T
A
£ +125∞C0.9 1.2 mA
INPUT CAPACITANCE
Differential C
IN
1.5 pF
Common Mode 10 pF
DYNAMIC PERFORMANCE
Slew Rate SR R
L
= 10 kW1V/ms
Overload Recovery Time 0.05 ms
Gain Bandwidth Product GBP 2 MHz
NOISE PERFORMANCE
Voltage Noise e
n
p-p 0.1 Hz to 10 Hz 0.5 mV p-p
Voltage Noise Density e
n
f = 1 kHz 22 nV/÷Hz
Current Noise Density i
n
f = 10 Hz 5 fA/÷Hz
Specifications subject to change without notice.
(VS = 2.7 V, VCM = 1.35 V, VO = 1.4 V, TA = 25ⴗC, unless otherwise noted.)
REV. A–4–
AD8628
ABSOLUTE MAXIMUM RATINGS
1
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V
Input Voltage . . . . . . . . . . . . . . . . GND – 0.3 V to V
S–
+ 0.3 V
Differential Input Voltage
2
. . . . . . . . . . . . . . . . . . . . . . . ±5.0 V
Output Short-Circuit Duration to GND . . . . . . . . . . Indefinite
Storage Temperature Range
R, RJ Packages . . . . . . . . . . . . . . . . . . . . . –65∞C to +150∞C
Operating Temperature Range . . . . . . . . . . –40∞C to +125∞C
Junction Temperature Range
R, RJ Packages . . . . . . . . . . . . . . . . . . . . . –65∞C to +150∞C
Lead Temperature Range (Soldering, 60 sec) . . . . . . . . 300∞C
1
Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those listed in the operational
sections of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
Differential input voltage is limited to ±5 V or the supply voltage, whichever is less.
Package Type ␪
JA
*␪
JC
Unit
5-Lead SOT-23 (RT-5) 230 146 ∞C/W
8-Lead SOIC (R) 158 43 ∞C/W
*␪
JA
is specified for worst-case conditions, i.e., ␪
JA
is specified for device soldered
in circuit board for surface-mount packages.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the AD8628 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.
ORDERING GUIDE
Temperature Package Package
Model Range Description Option Branding
AD8628ART-R2 –40∞C to +125∞C5-Lead SOT-23 RJ-5 AYA
AD8628ART-REEL7 –40∞C to +125∞C5-Lead SOT-23 RJ-5 AYA
AD8628AR –40∞C to +125∞C8-Lead SOIC R-8
AD8628AR-REEL –40∞C to +125∞C8-Lead SOIC R-8
AD8628AR-REEL7 –40∞C to +125∞C8-Lead SOIC R-8
REV. A
Typical Performance Characteristics–AD8628
–5–
INPUT OFFSET VOLTAGE (␮V)
–2.5 2.5
–1.5 –0.5 0.5 1.5
NUMBER OF AMPLIFIERS
180
160
0
80
60
40
20
140
100
120
V
S
= 2.7V
T
A
= 25ⴗC
TPC 1. Input Offset Voltage
Distribution at 2.7 V
INPUT OFFSET VOLTAGE (␮V)
–2.5 2 .5
–1.5 –0.5 0.5 1.5
NUMBER OF AMPLIFIERS
100
80
0
40
30
20
10
70
50
60
V
S
= 5V
V
CM
= 2.5V
T
A
= 25ⴗC
90
TPC 4. Input Offset Voltage
Distribution at 5 V
LOAD CURRENT (mA)
1
0.01
0.001
OUTPUT VOLTAGE (mV)
0.1 1 10
0.1
10
1k
100
0.0001 0.01
V
S
= 2.7V
SOURCE
SINK
TPC 7. Output Voltage to Supply
Rail vs. Load Current at 2.7 V
INPUT COMMON-MODE VOLTAGE (V)
INPUT BIAS CURRENT (pA)
60
001 6
23 45
+85ⴗC
40
30
20
10
+25ⴗC
–40ⴗC
50
VS = 5V
TPC 2. Input Bias Current vs. Input
Common-Mode Voltage at 5 V
TCVOS (nV/ⴗC)
NUMBER OF AMPLIFIERS
7
0
010
2468
6
5
4
3
2
1
V
S
= 5V
T
A
= –40ⴗC TO +125 ⴗC
TPC 5. Input Offset Voltage Drift
TEMPERATURE (ⴗC)
INPUT BIAS CURRENT (pA)
1,500
1,150
0
–50 –25 17502550 100 125 15075
800
450
100
VS = 5V
VCM = 2.5V
TA = –40ⴗC TO +150 ⴗC
TPC 8. Input Bias Current vs.
Temperature
INPUT COMMON-MODE VOLTAGE (V)
INPUT BIAS CURRENT (pA)
1,500
–1,500 01 6
23 45
150ⴗC
500
0
–500
–1,000
1,000
125ⴗC
V
S
= 5V
TPC 3. Input Bias Current vs. Input
Common-Mode Voltage at 5 V
LOAD CURRENT (mA)
1
0.01
0.001
OUTPUT VOLTAGE (mV)
0.1 1 10
0.1
10
1k
100
0.0001 0.01
V
S
= 5V
T
A
= 25ⴗC
SOURCE
SINK
TPC 6. Output Voltage to Supply
Rail vs. Load Current at 5 V
TEMPERATURE (ⴗC)
SUPPLY CURRENT (␮A)
1,250
1,000
0
–50 200050100 150
750
500
250
TA = 25ⴗC
5V
2.7V
TPC 9. Supply Current vs.
Temperature
REV. A–6–
AD8628
SUPPLY VOLTAGE (V)
SUPPLY CURRENT (␮A)
1,000
0
800
200
600
400
01 6
2345
TA = 25ⴗC
TPC 10. Supply Current vs.
Supply Voltage
FREQUENCY (Hz)
CLOSED-LOOP GAIN (dB)
50
–30
1k 10k 100k 1M 10M
30
20
10
0
–10
40
–20
70
60
A
V
= 100
A
V
= 10
A
V
= 1
V
S
= 2.7V
C
L
= 20pF
R
L
= 2k⍀
TPC 13. Closed-Loop Gain
vs. Frequency at 2.7 V
FREQUENCY (Hz)
OUTPUT IMPEDANCE (⍀)
100 1k 100M10k 100k 10M
300
270
0
240
210
180
150
120
90
60
30
VS = 5V
AV = 100
AV = 1
AV = 10
1M
TPC 16. Output Impedance
vs. Frequency at 5 V
FREQUENCY (Hz)
OPEN-LOOP GAIN (dB)
10k 100k 1M 10M
70
60
ⴚ30
50
40
30
20
10
0
ⴚ10
ⴚ20
0
45
90
135
180
225
PHASE SHIFT (Degrees)
V
S
= 2.7V
C
L
= 20pF
R
L
=
␾
M
= 52.1ⴗ
TPC 11. Open-Loop Gain
and Phase vs. Frequency
FREQUENCY (Hz)
CLOSED-LOOP GAIN (dB)
1k 10M10k 100k 1M
A
V
= 100
A
V
= 10
A
V
= 1
V
S
= 5V
C
L
= 20pF
R
L
= 2k⍀
50
–30
30
20
10
0
–10
40
–20
70
60
TPC 14. Closed-Loop Gain
vs. Frequency at 5 V
TIME (4␮s/DIV)
VOLTAGE (500mV/DIV)
0
0
0
00 0
00000000
0
0
0
0
0
0
V
S
= 1.35V
C
L
= 300pF
R
L
=
A
V
= 1
TPC 17. Large Signal Transient
Response at 2.7 V
FREQUENCY (Hz)
OPEN-LOOP GAIN (dB)
10k 100k 1M 10M
70
60
ⴚ30
50
40
30
20
10
0
ⴚ10
ⴚ20
45
90
135
180
225
0
PHASE SHIFT (Degrees)
VS = 5V
CL = 20pF
RL =
␾M = 52.1ⴗ
TPC 12. Open-Loop Gain
and Phase vs. Frequency
FREQUENCY (Hz)
OUTPUT IMPEDANCE (⍀)
100 1k 100M10k 100k 10M
300
270
0
240
210
180
150
120
90
60
30
VS = 2.7V
AV = 100
AV = 1
AV = 10
1M
TPC 15. Output Impedance
vs. Frequency at 2.7 V
TIME (5␮s/DIV)
VOLTAGE (1V/DIV)
0
0
0
00 0
00000000
0
0
0
0
0
0
V
S
= 2.5V
C
L
= 300pF
R
L
=
A
V
= 1
TPC 18. Large Signal Transient
Response at 5 V
REV. A
AD8628
–7–
TIME (4␮s/DIV)
VOLTAGE (50mV/DIV)
0
0
0
00 0
00000000
0
0
0
0
0
0
V
S
= 1.35V
C
L
= 50pF
R
L
=
A
V
= 1
TPC 19. Small Signal Transient
Response at 2.7 V
CAPACITIVE LOAD (pF)
OVERSHOOT (%)
110 1k100
80
0
70
60
50
40
30
20
10
OS+
OS–
VS = 2.5V
RL = 2k⍀
TA = 25ⴗC
TPC 22. Small Signal Overshoot
vs. Load Capacitance at 5 V
TIME (200␮s/DIV)
VOLTAGE (1V/DIV)
0
0
0
00 000000000
0
0
0
0
0
0
V
S
= 2.5V
V
IN
= 1kHz @ 3V p-p
C
L
= 0pF
R
L
= 10k⍀
A
V
= 1
TPC 25. No Phase Reversal
TIME (4␮s/DIV)
VOLTAGE (50mV/DIV)
0
0
0
00 0
00000000
0
0
0
0
0
0
VS = 2.5V
CL = 50pF
RL =
AV = 1
TPC 20. Small Signal Transient
Response at 5 V
TIME (2␮s/DIV)
VOLTAGE (V)
0
0
0
00 0
00000000
0
0
0
V
OUT
0
0
V
S
= 2.5V
A
V
= –50
R
L
= 10k⍀
C
L
= 0
CH1 = 50mV/DIV
CH2 = 1V/DIV
0V
V
IN
0V
TPC 23. Positive Overvoltage
Recovery
FREQUENCY (Hz)
CMRR (dB)
100 1k 10M10k 100k 1M
140
120
–60
100
80
60
40
20
0
–20
–40
V
S
= 2.7V
TPC 26. CMRR vs. Frequency at 2.7 V
CAPACITIVE LOAD (pF)
OVERSHOOT (%)
110 1k100
100
90
0
80
70
60
50
40
30
20
10
OS+
OS–
VS = 1.35V
RL = 2k⍀
TA = 25ⴗC
TPC 21. Small Signal Overshoot
vs. Load Capacitance at 2.7 V
TIME (10␮s/DIV)
VOLTAGE (V)
0
0
0
00 0
00000000
0
0
0
0
0
0
V
S
= 2.5V
A
V
= –50
R
L
= 10k⍀
C
L
= 0
CH1 = 50mV/DIV
CH2 = 1V/DIV
0V
V
IN
0V
V
OUT
TPC 24. Negative Overvoltage
Recovery
FREQUENCY (Hz)
100 1k 10M10k 100k 1M
V
S
= 5V
CMRR (dB)
140
120
–60
100
80
60
40
20
0
–20
–40
TPC 27. CMRR vs. Frequency at 5 V
REV. A–8–
AD8628
FREQUENCY (Hz)
100 1k 10M10k 100k 1M
VS = 1.35V
+PSRR
ⴚPSRR
PSRR (dB)
140
120
–60
100
80
60
40
20
0
–20
–40
TPC 28. PSRR vs. Frequency
FREQUENCY (Hz)
OUTPUT SWING (V p-p)
5.5
4.5
0
100 1k 1M
10k 100k
3.5
2.5
0.5
1.5
5.0
4.0
3.0
2.0
1.0
V
S
= 5V
R
L
= 10k⍀
T
A
= 25ⴗC
A
V
= 1
TPC 31. Maximum Output
Swing vs. Frequency at 5 V
VOLTAGE NOISE DENSITY (nV/ Hz)
FREQUENCY (kHz)
120
75
0
0 0.5 1.0 1.5 2.0 2.5
105
90
45
15
60
30
V
S
= 2.7V
NOISE AT 1kHz = 21.3nV
TPC 34. Voltage Noise Density at
2.7 V from 0 Hz to 2.5 kHz
FREQUENCY (Hz)
100 1k 10M10k 100k 1M
VS = 2.5V
+PSRR
ⴚPSRR
PSRR (dB)
140
120
–60
100
80
60
40
20
0
–20
–40
TPC 29. PSRR vs. Frequency
TIME (s)
VOLTAGE (␮V)
0.60
0.45
0.30
0.15
0
–0.15
–0.30
–0.45
–0.60
01 1023456789
V
S
= 2.7V
TPC 32. 0.1 Hz to 10 Hz
Noise at 2.7 V
VOLTAGE NOISE DENSITY (nV/ Hz)
FREQUENCY (kHz)
120
75
0
255101520
105
90
45
15
60
30
V
S
= 2.7V
NOISE AT 10kHz = 42.4nV
TPC 35. Voltage Noise Density
at 2.7 V from 0 Hz to 25 kHz
FREQUENCY (H)
OUTPUT SWING (V p-p)
3.0
2.5
0
100 1k 1M
10k 100k
2.0
1.5
0.5
1.0
VS = 2.7V
RL = 10k⍀
TA = 25ⴗC
AV = 1
TPC 30. Maximum Output
Swing vs. Frequency
TIME (␮s)
VOLTAGE (␮V)
VS = 5V
0.60
0.45
0.30
0.15
0
–0.15
–0.30
–0.45
–0.60
01 1023456789
TPC 33. 0.1 Hz to 10 Hz
Noise at 5 V
VOLTAGE NOISE DENSITY (nV/ Hz)
FREQUENCY (kHz)
120
75
0.5 2.51.0 1.5 2.0
105
90
45
15
60
30
VS = 5V
NOISE AT 1kHz = 22.1nV
TPC 36. Voltage Noise Density at
5 V from 0 Hz to 2.5 kHz
REV. A
AD8628
–9–
FREQUENCY – kHz
120
75
0
025
5101520
105
90
45
15
60
30
VOLTAGE NOISE DENSITY (nV/ Hz)
V
S
= 5V
NOISE AT 10kHz = 36.4nV
TPC 37. Voltage Noise Density at
5 V from 0 Hz to 25 kHz
TEMPERATURE (ⴗC)
OUTPUT SHORT-CIRCUIT CURRENT (mA)
150
–100
–50 –25 17502550 100 125 15075
VS = 2.7V
TA = –40ⴗC TO +150 ⴗC
–50
0
50
100
ISC–
ISC+
TPC 40. Output Short-Circuit
Current vs. Temperature
TEMPERATURE (ⴗC)
OUTPUT-TO-RAIL VOLTAGE (mV)
1k
100
0.10
10
1
–50 –25 17502550 100 125 15075
V
OL
– V
EE
@ 100k⍀
V
CC
– V
OH
@ 100k⍀
V
OL
– V
EE
@ 10k⍀
V
CC
– V
OH
@ 10k⍀
V
OL
– V
EE
@ 1k⍀
V
CC
– V
OH
@ 1k⍀
V
S
= 2.7V
TPC 43. Output-to-Rail Voltage
vs. Temperature
FREQUENCY – Hz
120
75
0
010
5
105
90
45
15
60
30
V
S
= 5V
VOLTAGE NOISE DENSITY (nV/ Hz)
TPC 38. Voltage Noise
TEMPERATURE (ⴗC)
OUTPUT SHORT-CIRCUIT CURRENT (mA)
150
–100
–50 –25 17502550 100 125 15075
VS = 5V
TA = –40ⴗC TO +150 ⴗC
–50
0
50
100
ISC–
ISC+
TPC 41. Output Short-Circuit
Current vs. Temperature
TEMPERATURE (ⴗC)
POWER SUPPLY REJECTION (dB)
150
50
–50 –25 0 25 50 100 12575
VS = 2.7V TO 5V
TA = –40ⴗC TO +125 ⴗC
60
70
80
90
100
100
120
130
140
TPC 39. Power Supply Rejec-
tion vs. Temperature
TEMPERATURE (ⴗC)
OUTPUT-TO-RAIL VOLTAGE (mV)
1k
100
0.10
10
1
–50 –25 17502550 100 125 15075
V
OL
– V
EE
@ 100k⍀
V
CC
– V
OH
@ 100k⍀
V
OL
– V
EE
@ 10k⍀
V
CC
– V
OH
@ 10k⍀
V
OL
– V
EE
@ 1k⍀
V
CC
– V
OH
@ 1k⍀
V
S
= 5V
TPC 42. Output-to-Rail Voltage
vs. Temperature
REV. A–10–
AD8628
FUNCTIONAL DESCRIPTION
The AD8628 is a single-supply, ultrahigh precision rail-to-rail
input and output operational amplifier. The typical offset voltage
of less than 1 mV allows this amplifier to be easily configured for
high gains without risk of excessive output voltage errors. The
extremely small temperature drift of 2 nV/∞C ensures a minimum
of offset voltage error over its entire temperature range of –40∞C
to +125∞C, making the AD8628 amplifier ideal for a variety of
sensitive measurement applications in harsh operating environ-
ments. The AD8628 achieves a high degree of precision through
a patented combination of auto-zeroing and chopping. This
unique topology allows the AD8628 to maintain its low offset
voltage over a wide temperature range and over its operating
lifetime. AD8628 also optimizes the noise and bandwidth over
previous generations of auto-zero amplifiers, offering the lowest
voltage noise of any auto-zero amplifier by more than 50%.
Previous designs used either auto-zeroing or chopping to add
precision to the specifications of an amplifier. Auto-zeroing
results in low noise energy at the auto-zeroing frequency at the
expense of higher low frequency noise due to aliasing of wideband
noise into the auto-zeroed frequency band. Chopping results in
lower low frequency noise at the expense of larger noise energy
at the chopping frequency. AD8628 uses both auto-zeroing and
chopping in a patented ping-pong arrangement to obtain lower
low frequency noise together with lower energy at the chopping
and auto-zeroing frequencies, maximizing the signal-to-noise
ratio (SNR) for the majority of applications without the need
for additional filtering. The relatively high clock frequency of
15 kHz simplifies filter requirements for a wide, useful, noise-
free bandwidth.
AD8628 is one of the few auto-zero amplifiers offered in the 5-lead
SOT-23 package. It greatly improves the ac parameters of the
previous auto-zero amplifiers. It has low noise over a relatively wide
bandwidth (0 Hz to 10 kHz) and can be used where the highest
dc precision is required. In systems with signal bandwidths up to
5 kHz to 10 kHz, the AD8628 provides true 16-bit accuracy
making it the best choice for very high resolution systems.
1/f Noise
1/f noise, also known as “pink noise,” is a major contributor of
errors in dc-coupled measurements. This 1/f noise error term
can be in the range of several mV or more, and, when amplified
with the closed-loop gain of the circuit, can show up as a large
output offset. For example, when an amplifier with a 5 mV p-p 1/f
noise is configured for a gain of 1,000, its output will have 5 mV
of error due to the 1/f noise. But AD8628 eliminates 1/f noise
internally and therefore greatly reduces output errors. Here is how
it works: 1/f noise appears as a slowly varying offset to AD8628
inputs. Auto-zeroing corrects any dc or low frequency offset, thus
the 1/f noise component is essentially removed, leaving AD8628
free of 1/f noise.
FREQUENCY (kHz)
120
105
00124
VOLTAG E NOISE DENSITY (nV/ Hz)
60
45
30
15
90
75
26810
AD8628
(19.4nV/ Hz)
LMC2001
(31.1nV/ Hz)
LT C2050
(89.7nV/ Hz)
MK AT 1kHz FOR ALL 3 GRAPHS
Figure 1. Noise Spectral Density of AD8628 vs.
Competition
One of the biggest advantages that AD8628 brings to systems
applications over competitive auto-zero amplifiers is its very low
noise. The comparison shown in Figure 1 indicates an input-
referred noise density of 19.4 nV/÷Hz at 1 kHz for AD8628 that
is much better than the LTC2050 and LMC2001. The noise is
flat from dc to 1.5 kHz, slowly increasing up to 20 kHz. The lower
noise at low frequency is desirable where auto-zero amplifiers
are widely used.
REV. A
AD8628
–11–
AD8628 Peak-to-Peak Noise vs. Competition
Because of the ping-pong action between auto-zeroing and
chopping, the peak-to-peak noise of the AD8628 is much lower
than its competition. Figures 2 and 3 show this comparison.
TIME (1s/DIV)
0
0
0
000
VOLTAG E (0.5␮V/DIV)
00000000
0
0
0
0
0
0
e
n
p-p = 0.5␮V
BW = 0.1Hz to 10Hz
Figure 2. AD8628 Peak-to-Peak Noise
TIME (1s/DIV)
0
0
0
000
VOLTAG E (0.5␮V/DIV)
00000000
0
0
0
0
0
0
e
n
p-p = 2.3␮V
BW = 0.1Hz to 10Hz
Figure 3. LTC2050 Peak-to-Peak Noise
Noise Behavior with First Order Low-Pass Filter
AD8628 was simulated as a low-pass filter and then configured
as shown in Figure 4. The behavior of the AD8628 matches the
simulated data. It was verified that noise is rolled off by first
order filtering.
IN
OUT
470pF100k⍀
1k⍀
Figure 4. Test Circuit: First Order Low-Pass
Filter—x101 Gain and 3 kHz Corner Frequency
FREQUENCY (Hz)
10010 20 30 40 50 60 70 80 90
45
20
NOISE (dB)
30
10
5
25
15
40
35
50
Figure 5a. Simulation Transfer Function of Test Circuit
FREQUENCY (kHz)
45
20
NOISE (dB)
30
10
5
25
15
40
35
50
Figure 5b. Actual Transfer Function of Test Circuit
Measured noise spectrum of test circuit showing noise between
5 kHz and 45 kHz is successfully rolled off by first order filter.
Total Integrated Input-Referred Noise for First Order Filter
(AD8628 vs. Competition)
3dB FILTER BANDWIDTH (Hz)
10
1
0.110 10k100
RMS NOISE (␮V)
1k
LT C2050
AD8551 AD8628
Figure 6. 3 dB Filter Bandwidth in Hz
For a first order filter, the total integrated noise from the AD8628
is lower than the LTC2050.
REV. A–12–
AD8628
Input Overvoltage Protection
Although the AD8628 is a rail-to-rail input amplifier, care should
be taken to ensure that the potential difference between the inputs
does not exceed the supply voltage. Under normal negative feed-
back operating conditions, the amplifier will correct its output to
ensure the two inputs are at the same voltage. However, if either
input exceeds either supply rail by more than 0.3 V, large currents will
begin to flow through the ESD protection diodes in the amplifier.
These diodes are connected between the inputs and each supply
rail to protect the input transistors against an electrostatic dis-
charge event and are normally reverse biased. However, if the
input voltage exceeds the supply voltage, these ESD diodes will
become forward biased. Without current limiting, excessive
amounts of current could flow through these diodes, causing
permanent damage to the device. If inputs are subject to
overvoltage, appropriate series resistors should be inserted to limit
the diode current to less than 5 mA maximum.
Output Phase Reversal
Output phase reversal occurs in some amplifiers when the input
common-mode voltage range is exceeded. As common-mode
voltage is moved outside of the common-mode range, the outputs
of these amplifiers will suddenly jump in the opposite direction
to the supply rail. This is the result of the differential input pair
shutting down, causing a radical shifting of internal voltages that
results in the erratic output behavior. The AD8628 amplifier
has been carefully designed to prevent any output phase reversal,
provided both inputs are maintained within the supply voltages.
If one or both inputs could exceed either supply voltage, a resistor
should be placed in series with the input to limit the current to less
than 5 mA. This will ensure the output will not reverse its phase.
Overload Recovery Time
Many auto-zero amplifiers are plagued by long overload recovery
time, often in milliseconds, due to the complicated settling
behavior of the internal nulling loops after saturation of the
outputs. AD8628 has been designed so that internal settling
occurs within two clock cycles after output saturation happens.
This results in a much shorter recovery time, less than 10 ms, when
compared to other auto-zero amplifiers. The wide bandwidth of
the AD8628 enhances performance when it is used to drive loads
that inject transients into the outputs. This is a common
situation when an amplifier is used to drive the input of switched
capacitor ADCs.
TIME (500␮s/DIV)
0
0
0
000
VOLTAG E ( V)
00000000
0
0
0
0
0
0
VIN
0V
0V
VOUT
CH 1 = 50mV/DIV
CH 2 = 1V/DIV
AV = –50
Figure 7. Positive Input Overload Recovery for AD8628
TIME (500␮s/DIV)
0
0
0
000
VOLTAG E ( V)
00000000
0
0
0
0
0
0
VIN
0V
0V
VOUT
CH 1 = 50mV/DIV
CH 2 = 1V/DIV
AV = –50
Figure 8. Positive Input Overload Recovery for LTC2050
TIME (500␮s/DIV)
0
0
0
000
VOLTAG E ( V )
00000000
0
0
0
0
0
0
VIN
0V
0V
VOUT
CH 1 = 50mV/DIV
CH 2 = 1V/DIV
AV = –50
Figure 9. Positive Input Overload Recovery for LMC2001
REV. A
AD8628
–13–
TIME (500␮s/DIV)
0
0
0
000
VOLTAG E ( V )
00000000
0
0
0
0
0
0
VIN
0V
0V
VOUT
CH 1 = 50mV/DIV
CH 2 = 1V/DIV
AV = –50
Figure 10. Negative Input Overload Recovery for AD8628
TIME (500␮s/DIV)
0
0
0
000
VOLTAG E ( V )
00000000
0
0
0
0
0
0
VIN
0V
0V
VOUT
CH 1 = 50mV/DIV
CH 2 = 1V/DIV
AV = –50
Figure 11. Negative Input Overload Recovery for LTC2050
TIME (500␮s/DIV)
0
0
000
VOLTAG E ( V)
00000000
0
0
0
0
0
0
VIN
0V
0V
VOUT
CH 1 = 50mV/DIV
CH 2 = 1V/DIV
AV = –50
0
Figure 12. Negative Input Overload Recovery for LMC2001
The results shown in Figures 7–12 are summarized in Table I.
Table I. Overload Recovery Time
Product Type Positive Overload Negative Overload
Recovery Recovery (␮s) Recovery (␮s)
AD8628 6 9
LTC2050 650 25,000
LMC2001 40,000 35,000
Infrared Sensors
Infrared (IR) sensors, particularly thermopiles, are increasingly
being used in temperature measurement for applications as wide-
ranging as automotive climate controls, human ear thermometers,
home insulation analysis, and automotive repair diagnostics.
The relatively small output signal of the sensor demands high
gain with very low offset voltage and drift to avoid dc errors. If
interstage ac coupling is used (Figure 13), low offset and drift
prevents the input amplifier’s output from drifting close to satu-
ration. The low input bias currents generate minimal errors from
the sensor’s output impedance. As with pressure sensors, the
very low amplifier drift with time and temperature eliminates
additional errors once the temperature measurement has been
calibrated. The low 1/f noise improves SNR for dc measurements
taken over periods often exceeding 1/5 second. Figure 15 shows a
circuit that can amplify ac signals from 100 mV to 300 mV up to
the 1 V to 3 V level, gain of 10,000 for accurate A/D conversion.
5V
100k⍀10k⍀
5V
100␮V – 300␮V
100⍀
TO BIAS
VOLTAG E
10k⍀
fC 1.6Hz
IR
DETECTOR
100k⍀
10␮F
AD8628
AD8628
Figure 13. Preamplifier for Thermopile
REV. A–14–
AD8628
Precision Current Shunts
A precision shunt current sensor benefits from the unique attributes
of auto-zero amplifiers when used in a differencing configuration
(Figure 14). Shunt current sensors are used in precision current
sources for feedback control systems. They are also used in a
variety of other applications, including battery fuel gauging, laser
diode power measurement and control, torque feedback con-
trols in electric power steering, and precision power metering.
R
S
0.1⍀
SUPPLY IR
L
100⍀100k⍀
C
5V
100⍀100k⍀
C
e = 1,000 R
S
I
100mV/mA
AD8628
Figure 14. Low-Side Current Sensing
In such applications, it is desirable to use a shunt with very low
resistance to minimize the series voltage drop; this minimizes
wasted power and allows the measurement of high currents with-
out saving power. A typical shunt might be 0.1 W. At measured
current values of 1 A, the shunt’s output signal is hundreds of
millivolts, or even volts, and amplifier error sources are not
critical. However, at low measured current values in the 1 mA
range, the 100 mV output voltage of the shunt demands a very
low offset voltage and drift to maintain absolute accuracy. Low
input bias currents are also needed, so that “injected” bias current
does not become a significant percentage of the measured current.
High open-loop gain, CMRR, and PSRR all help to maintain
the overall circuit accuracy. As long as the rate of change of the
current is not too fast, an auto-zero amplifier can be used with
excellent results.
Output Amplifier for High Precision DACs
AD8628 is used as an output amplifier for a 16-bit high precision
DAC in unipolar configuration. In this case, the selected op amp
needs to have very low offset voltage (the DAC LSB is 38 mV when
operated with a 2.5 V reference) to eliminate the need for output
offset trims. Input bias current (typically a few tens of pico amp)
must also be very low since it generates an additional zero code
error when multiplied by the DAC output impedance (approxi-
mately 6 kW). Rail-to-rail input and output provide full-scale
output with very little error. Output impedance of the DAC is
constant and code-independent, but the high input impedance
of the AD8628 minimizes gain errors. The amplifier’s wide band-
width also serves well in this case. The amplifier with settling time
of 1 ms adds another time constant to the system, increasing the
settling time of the output. The settling time of the AD5541 is
1ms. The combined settling time is approximately 1.4 ms, as can be
derived from the equation:
t TOTAL t DAC t AD
SSS
()
=
()
+
()
22
8628
10␮F
REF(REF*)V
DD
REFS*
CS
DIN
SCLK
LDAC*DGND AGND
OUT UNIPOLAR
OUTPUT
AD8628
*AD5542 ONLY
SERIAL
INTERFACE
0.1␮F
5V
0.1␮F
2.5V
AD5541/AD5542
Figure 15. AD8628 Used as an Output Amplifier
REV. A
AD8628
–15–
OUTLINE DIMENSIONS
5-Lead Small Outline Transistor Package [SOT-23]
(RT-5)
Dimensions shown in millimeters
PIN 1
1.60 BSC 2.80 BSC
1.90
BSC
0.95 BSC
1 3
4 5
2
0.22
0.08
10ⴗ
5ⴗ
0ⴗ
0.50
0.35
0.15 MAX SEATING
PLANE
1.45 MAX
1.30
1.15
0.90
2.90 BSC
0.60
0.45
0.30
COMPLIANT TO JEDEC STANDARDS MO-178AA
8-Lead Standard Small Outline Package [SOIC]
(R-8)
Dimensions shown in millimeters and (inches)
0.25 (0.0098)
0.17 (0.0067)
1.27 (0.0500)
0.40 (0.0157)
0.50 (0.0196)
0.25 (0.0099) ⴛ 45ⴗ
8ⴗ
0ⴗ
1.75 (0.0688)
1.35 (0.0532)
SEATING
PLANE
0.25 (0.0098)
0.10 (0.0040)
85
41
5.00 (0.1968)
4.80 (0.1890)
4.00 (0.1574)
3.80 (0.1497)
1.27 (0.0500)
BSC
6.20 (0.2440)
5.80 (0.2284)
0.51 (0.0201)
0.31 (0.0122)
COPLANARITY
0.10
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MS-012AA
REV. A
C02735–0–6/03(A)
–16–
AD8628
REVISION HISTORY
Location Page
6/03—Data Sheet changed from Rev. 0 to Rev. A
Changes to SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Changes to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Change to FUNCTIONAL DESCRIPTION section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15